Power-conversion engine

ABSTRACT

The disclosure provides a power converter and method for controlling same, comprising a plurality of switch elements, an inductive reactor, and at least two ports for the movement of electrical energy. Any energy-moving port may be made unipolar, bidirectional, bipolar, or bidirectionally bipolar. Ports may be equipped with sensing circuitry to allow the converter output to be controlled responsively to an input signal. The disclosure may be configured to be used in many ways, for example, as a power-supply, as an amplifier, or as a frequency converter. The disclosure may comprise energy predictive calculating means to obtain excellent transient response to line and load variations. The disclosure may also include a switch to create a low impedance path around the inductor to allow current to recirculate through the inductor when it is not needed at any of the ports.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. application Ser. No.12/840,436 filed Jul. 21, 2010, incorporated herein by reference, whichis itself a divisional of U.S. application Ser. No. 11/682,169 filedMar. 5, 2007, incorporated herein by reference, which claims the benefitof U.S. Provisional Application No. 60/823,564 filed on Aug. 25, 2006,which is incorporated herein by reference.

BACKGROUND

Electrical power conversion devices are well known in the prior art andinclude power supplies, amplifiers, converters (AC-DC, AC-AC, DC-DC andDC-AC), generators and drives.

Some prior-art supplies are switched forward-converters that depend ontransformer turns-ratio to accommodate differing input and outputvoltage in which energy stored in an inductive field is a mereby-product of operation.

Other very common power converters related to the present disclosure areswitched supplies operating in either the well-known buck or flybackmodes. A few supplies such as that of U.S. Pat. No. 6,275,016 arecapable of both buck and boost operation. Some flyback supplies are usedto provide polarity inversion between input and output.

Sometimes multiple supplies are used to provide bidirectional flow ofpower, but recently integrated bidirectional supplies have beendeveloped. The supply of U.S. Pat. No. 5,734,258 is bidirectional andoperates in both buck and boost modes. U.S. Pat. No. 7,046,525 describesa bidirectional flyback mode supply. U.S. Pat. Nos. 6,894,461 and5,196,995 describe bidirectional supplies.

Some supplies like U.S. Pat. No. 6,894,461 have a plurality of energyports.

Less common are switched mode amplifiers with inductive energy-storageelements. The amplifier of U.S. Pat. No. 4,186,437 uses a plurality ofinductors to obtain bipolar operation. The amplifier of U.S. Pat. No.7,030,694 uses a plurality of DC-DC converters, complementarilycontrolled, to achieve bipolar operation.

The power converter of U.S. Pat. No. 5,196,995 is bidirectional andbipolar, but is bidirectional only when inverting polarity.

It is not known in the prior art to provide an integrated switched-modepower-converter that is bidirectionally bipolar.

SUMMARY

The present disclosure provides a switched mode power converter andmethod for controlling same, comprising a plurality of switch elements,an inductive reactor, and at least two ports for the movement ofelectrical energy. The arrangement of switches of this disclosureenables an inductive reactor, such as an inductor or transformer, toparticipate in power conversion between ports. Such conversion mayemploy the buck or flyback modes, or both. Any energy-moving port may bemade unipolar, bidirectional, bipolar, or bidirectionally bipolar. Portsmay or may not be galvanically isolated by using a transformer as aninductive reactor. Ports may be equipped with sensing circuitry to allowthe converter output to be controlled responsively to an input signal.The disclosure may be configured to be used in many ways, for example,as a power-supply, as an amplifier, or as a frequency converter. Thedisclosure may comprise energy predictive calculating means to obtainexcellent transient response to line and load variations. The disclosuremay also comprise means to enable it to adapt to changes of internaland/or external reactive components. The disclosure may also include aswitch to create a low impedance path around the inductor to allowcurrent to recirculate through the inductor when it is not needed at anyof the ports.

In one embodiment, the disclosure comprises power input and outputports, an inductive reactor for energy storage, switches for connectingthe ports to the inductive reactor, a sensor generating a feedbacksignal responsive to the voltage or current at the output port, areference signal and a control circuit. The control circuit controls theswitches in response to the feedback signal and reference signal so thatthe polarity of the input and output ports can be switched betweeninverting and non-inverting and such that for any polarity energy mayflow from the input port to the output port or from the output port tothe input port. In one preferred embodiment of this implementation ofthe disclosure the amount of energy placed into the inductive reactorduring charging of the inductive reactor is controlled basedapproximately on upon the per-chopping-cycle load energy requirement atthe output port.

In another similar embodiment of the disclosure one port is bipolar andanother port is unipolar and the converter may be inverting ornon-inverting. The inductive energy element is an inductor and it isdirectly coupled to the ports through the switches. The control circuitmaintains a desired relationship between the voltage or current at theoutput port and the reference signal.

In one method of the disclosure the converter is comprised of aninductive reactor, at least two power-moving ports, a reference signaland a number of switches. The method involves setting the switches inresponse to the sign and magnitude of the inductive reactor current tomaintain the voltage or current at one of the ports in a desiredmathematical relationship with the reference signal. In one embodimentof this method the method involves selecting a mode of converteroperation based up the magnitude and sign of the reference signal, themagnitude and sign of the power-moving port voltages and the magnitudeand sign of the inductive reactor current. Once the mode is selected,the switches are set to effect this mode.

In another embodiment the disclosure comprises a method of operating aswitched mode power supply with a plurality of modes. The modes includeat least an inductive reactor energizing mode and energy transfer modewherein energy is transferred to at least one power-moving port. Thesupply is operated so that more than one mode may occur during anysingle chopping cycle. Preferably, this method also allows a mode tooverlap chopping cycles so that the mode at the end of a given choppingcycle is the mode at the beginning of the next chopping cycle.

In another embodiment of the disclosure the converter is configured as aswitched mode flyback converter and has circuitry for calculatingpredicted pedestal current in the inductor based upon a feedback signalresponsive the voltage or current at a port and the reference signal.The control circuitry commutates the switches in response to thepredicted pedestal current and the feedback signal such that a desiredmathematical relationship is maintained between the output port and areference signal.

In another embodiment of the disclosure the converter is configured as aswitched mode power converter comprising an inductive reactor, first andsecond switches and an auxiliary energy source (such as a reactor,capacitor, inductor, battery, generator or power supply). The method forcontrolling this embodiment is comprised of predicting within a choppingcycle the inductive pedestal current at the end of the present cycle,control the first switch such that the per-cycle pedestal current changeis limited, and controlling the second switch in response to the currentpedestal predictions to move energy between the inductive reactor andthe auxiliary energy source.

In another embodiment of the disclosure the converter is configured tooperate in bipolar, bidirectional, inverting or flyback modes and iscomprised of a power input port, switches, an inductor and a poweroutput port. The converter also has an alternate path that allows energyto flow from the inductor other than to the output port. The alternatepath could be a low impedance or short circuit around the inductorallowing current to recirculate through the inductor, a path from theinductor back to the input port, or a path to a third port. The methodof operating this converter is comprised of transferring energy from thepower input port to the inductor to energize the inductor, transferringpower from the inductor to the power output port and controllablyconnecting the inductor to the power output port or the alternate pathin order to control the power at the power output port. This method maybe adapted so that the output voltage is lower than the input voltage byshortening the period of the flyback mode.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 depicts a bidirectionally bipolar switched mode power converteraccording to this disclosure.

FIG. 2 depicts possible directions and polarity of energy movementbetween two energy moving ports of the power converter of thisdisclosure.

FIG. 3 depicts energy balancing functions comprised in anenergy-predicting calculator that may be embedded within thepower-conversion engine according to this disclosure.

FIGS. 4A-4J depict different modes of a switching arrangement for thebidirectionally bipolar transfer of energy between non-isolated portsusing an inductor and six switches.

FIGS. 5A-5D show different portions of a flowchart depicting the logicto implement a bipolar, bidirectional, buck-boost, energy-balancing,synchronous power-conversion amplifier using the converter of FIGS.4A-4J.

FIGS. 6A-6C depict different modes of a bidirectional switch suitablefor use in this disclosure.

FIG. 7 depicts current and voltage related to the switch shown in FIGS.6A-6C.

FIG. 8 shows a schematic of a practical switch and driver arrangementfor the switches of this disclosure.

FIGS. 9A and 9B are timing diagrams showing the mode of the powerconverter of FIGS. 5A-5D as driven by the flow chart of FIG. 7 forpositive and negative PA voltages, respectively.

FIG. 10 shows an energy-balancing calculator according to thisdisclosure.

FIGS. 11A-11H show certain useful states of the disclosure.

FIG. 12 represents a flyback amplifier embodying both CCM stability andenergy balancing according to this disclosure.

FIG. 13 shows the performance of the amplifier in FIG. 12.

FIG. 14 also shows the performance of the amplifier in FIG. 12.

FIG. 15 also shows the performance of the amplifier in FIG. 12.

FIG. 16 shows a flyback converter of the current disclosure with currentrecirculation.

FIG. 17 shows the flyback converter of FIG. 16 with adaptation added.

FIG. 18 shows the voltage and current waveforms for the converter ofFIG. 17.

FIG. 19 is an enlargement of a portion of FIG. 18 showing recirculationaction during a negative load transition.

FIGS. 20A and 20B show a four-switch bidirectional flyback amplifierwith energy balance and continuous mode operation, respectively.

FIG. 21 shows the voltage and current waveforms for the converter ofFIGS. 20A and 20B.

FIG. 22 shows a state decoder used in the implementation shown in FIG.24.

FIG. 23 shows a schematic of the switch blocks used in the switches ofthe implementation shown in FIG. 24.

FIG. 24 shows a five switch bidirectional flyback amplifier withinductive energy storage.

FIG. 25 shows the voltage and current waveforms for the converter ofFIG. 24.

FIG. 26 shows additional performance of the converter in FIG. 24.

FIG. 27 shows additional performance of the converter in FIG. 24.

DETAILED DESCRIPTION

To avoid confusion, let the terms defined hereinbelow be used todescribe this disclosure:

Definitions

1. “PORT” means a galvanically related pair or group of terminalsdisposed to comprise a portion of a circuit to be externally attached tothis disclosure. Most ports are energy-moving ports. There may be acontrol port, comprising one or more connection-pairs to externalcircuits, for the purpose of controlling the operation of thispower-conversion engine.

2. An “ACTIVE” port means one that maintains a voltage or currentresponsive to energy calculations. An active port usually comprisespresently-working sense means in addition to energy-moving means. A“PASSIVE” port defaults to conditions set by energy present co-actingwith an external circuit attached thereto.

3. “FORWARD” (FWD) indicate energy transfer from a passive port to anactive port, whilst “REVERSE” (REV) indicates transfer from an activeport to a passive port.

4. “DISSIPATOR” indicates that a port is consuming energy being suppliedby an external circuit, whilst “GENERATOR” indicates that a port issupplying energy to and external circuit. Note that it is a practicalmode for an active port to be a dissipator.

5. “NON-INVERSION” (NON) indicates no polarity inversion between adissipator port and a generator port, whilst “INVERSION” (INV) indicatespolarity inversion between such ports. “Inverting” means that polarityis switched between the ports (for example if the dissipater port has apositive voltage the generator port has a negative voltage)”

6. “ENERGIZE” (EN) indicates the act of increasing inductive energy atthe expense of an external circuit attached to a dissipator port, whilst“TRANSFER” (TR) indicates the act of decreasing inductive energy througha generator port.

7. “CHARGE” indicates the act of increasing, and “DISCHARGE” act ofdecreasing, the energy of a capacitor or battery.

8. “BUCK” indicates the well-known buck mode of power conversion,conventionally employed when the turns-ratio-adjusted voltage across adissipator port exceeds that across an associated generator port. “FLY”indicates the well-known flyback, or boost, mode of power conversion,usually employed when the turns-ratio-adjusted voltage across agenerator port exceeds that across an associated dissipator port.

9. “balf” and “balfb” indicate respectively attainment of predictedenergy balance during forward and forward-buck energizing times. Theexpressions “until balf” and “until balfb” indicate that inductiveenergizing is to continue until such a balance is attained.

10. “I sub th” (Ith) indicates the attainment of a low-currentthreshold, representing a state of inconsequential energy available fromall inductive reactors substantially involved in energy transfer betweenports. The expression “until Ith” indicates that an energy transfer isto continue until this state is attained.

11. The “INDUCTOR” means a two terminal inductive reactor, whilst theterm “INDUCTIVE REACTOR” includes both inductors and other inductivereactors, such as transformers, having more than two terminals connectedto one or more windings.

12. “Bipolar” refers to a converter that can be either inverting ornon-inverting between a given two ports. Bipolar also refers to a portthat is capable of energizing or transferring both polarities ofvoltage.

13. “Bidirectional” means a) with respect to a port, that port can actas a dissipater and a generator; b) with respect to a converter that fora given two ports, power can flow into a first port and out of a secondport, and at a different time, out of the first port and into the secondport.

14. “Estimating” means determining the approximate voltage or current ata given location in a circuit or port either by measuring orcalculation, and also includes measuring or calculating a valueproportional to said voltage or current.

15. “Recirculate” or “recirculation” refers to creating a low impedancepath or short circuit around an inductor such that current circulatesout of the inductor, across the low impedance path or short circuit, andback into the inductor, preferably with very low losses.

16. “Direct-Coupled” means that energy moves between ports and throughthe inductive reactor as current flowing in a complete galvanic circuit.For example, ports that are coupled through transformers or capacitorswould not be direct coupled.

FIG. 1 shows a power converter 100 using six switches S1 through S6 toconnect two ports PA and PB, to inductor L (used for energy storage).This figure depicts the most general embodiment the present disclosure.It should be noted that ports PA and PB may be identical but areseparately referenced to emphasize that each may be made capable of thefull performance to be described below in FIG. 2 and that the ports ofconverter 100 are not being exchanged to obtain such comprehensivebehavior. Shown accompanying, and potentially connected to, each of thetwo ports PA and PB of the converter 100 are three energy sources VAC,VDC−, and VDC+, and a load LD. Using the topology shown therein, alloctants of energy transfer shown in FIG. 2 below may be accomplished atwill using the novel switch control circuitry described hereinbelow. Anyof the sources or loads shown near ports PA and PB may attach to theseports to effect any or all the energy transfer octants of FIG. 2, withtwo fundamental limitations.

The first limitation is that if energy-devoid loads be connected to theports PA and PB, and no other port is present, only such energy as isheld within the inductor L can be transferred to a port. The secondsimilar limitation is that the converter 100 is usable for AC-ACconversion only if fitted with an additional port connected to an energystorage device (such as a capacitor), or if the input and output ACwaves are substantially of the same frequency and remain in phase. Thenecessary energy storage device may alternatively be placed at PA or PBor at another port.

While a similar six switch topology is known in the prior art, no suchprior art converter has been equipped with control circuitry to enable asingle embodiment thereof to transfer energy from any of the sourcesshown in this figure to any load shown therein in any direction. Usingthe control circuitry of this disclosure shown below, the converter 100can not only transfer energy from any of the sources shown, but alsoadjusts its operation to accommodate independent changes of any suchsources from AC to DC of either polarity, or even from being energysources to being loads.

Less comprehensive sets of behavior than the set shown by all of theoctants of FIG. 2 may also be performed by converter 100 includingbidirectional, bipolar, unidirectional, and unipolar energy transfers.Many of the embodiments discussed below requiring less comprehensivebehavior may be practiced with fewer switches than are shown in thisfigure, or with diodes replacing some switches.

FIG. 2 depicts eight energy movement octants, I through VIII, betweentwo ports, PA and PB. Each octant shows an instance of converter 100performing a unique polarity and direction of energy movement. Theseoctants show the elemental aspects of bidirectional and bipolar energytransfer. In the first four octants port PA acts as a dissipator port(energy flowing into converter 100) and PB acts as a generator port(energy flowing out of converter 100), whereas in the last four octantstheir roles are reversed. In octants I, II, V, and VI, the dissipatorport sinks energy from a positive source, whilst in octants III, IV,VII, and VIII the dissipator port sinks energy from a negative source,and in all the generator port sources energy to the load LD. In alleven-numbered octants the ports are co-polar, but in all odd-numberedoctants the relative polarities of ports are inverted. Thus we see thatsince either port may have either polarity, energy conversion may befrom DC-DC, or DC-AC in either direction subject to energy limitationscited above. Bidirectional bipolar energy movement is supported. Noknown existing integrated switched-mode power-converter embodies all ofthese octants. These octants should not be confused with the “fourquadrants” of some bipolar converters that usually indicate bipolarityof both voltage and current; clearly bidirectional current must flow inthese octants. The switches S1 through S6 of converter 100 must becontrolled in an ordered manner to effect the multiplicity of energymovements shown. Each octant may in fact require multiple orderedsettings of these switches to perform its energy movement. To this endmany switch settings, or modes, subdivide the octants shown in thisfigure, and are shown below in detail in FIGS. 4A-4J.

For this disclosure, the port being used as an output is called a“generator” port and may be fitted with output sensing circuit, whilstthe port being used as an input is called a “dissipator” port. Theenergy sources shown and resistor LD shown as a load in FIGS. 1 and 2are generic and represent any electrical energy source or sink. Suchsources as DC power supplies, AC mains, and batteries are usual. Suchloads as electronic circuits are typical. Various motors may also beloads, for example a DC-AC embodiment according to this figure could beused for speed control of a synchronous motor. The load LD is notlimited to resistive but may be capacitive, inductive or any combinationthereof. The present disclosure of this figure might be especiallyuseful for efficient cyclic driving of piezo-electric motors, inasmuchas much energy stored in the capacitance thereof to cause a motion canbe returned to a power source upon reversal of said motion.

FIG. 3 depicts a multi-port embodiment of the converter 100 and possibleenergy balancing functions comprised within in an energy-predictingcalculator 150 that may be embedded within this power-conversion engine.The energy prediction methodology described herein is one novel aspectof the present disclosure. Energy balance according to this disclosureis attained when the energy predicted to be extractable from aninductive reactance substantially equals that predicted to be requiredby a load plus any losses, plus any deficit or minus any surplus ofenergy required to bring any capacitance to a desired voltage at a giventime. The calculator 150 may be implemented using well-known analog ordigital techniques.

When this disclosure is practiced in an energy balancing mode, it isfundamental that an inductive reactor within the engine be cyclicallyenergized with an amount of energy supply that is predicted, upontransfer, to meet a demand until said reactor be re-energized and a newtransfer accomplished. To this end, a term “KEL” within calculator 150represents energy stored within said reactor, and sometimes representsonly the extractable portion of that energy. Circuitry KEL for providingKEL may comprise circuits responsive inductive current, but may alsocomprise a circuit responsive to the volt-time product having beenapplied to said reactor, or one or more sensors responsive to themagnetic field of said reactor.

Another term “KEld” of energy balance according to this disclosurerepresents the predicted energy demand of a load during a time ofprediction. Circuitry KEld for generating this term comprises circuitryresponsive to three quantities. A current sub-term may be responsive toinductive current, but may be derived from the voltage droop across afilter capacitor during a particular time period, or may even bedirectly measured at a load. A voltage term is usually obtained bysensing voltage at an active port. A time term is usually generated bytiming means integral to the energy-predicting calculator 150. Theproduct of current, voltage, and time yields the term here KEld.

In many instances a load will be reactive, and usually it will becapacitive. According to this disclosure it is advantageous to use theterm “KEC” to represent the energy that will be required to bring thevoltage across any filter capacitance internal to and attached to anactive port to a desired voltage a given time or point in a cycle.Circuitry KEC to generate KEC is usually responsive to port-voltagesensing and to a desired input signal or a voltage reference. Since KECis predictively rather than historically calculated, its use accordingto this disclosure in closing the feedback loop eliminates the pole thatplagues the control loops of prior-art amplifiers and power supplies,yielding unprecedented transient response.

KEL relates to inductive reactor value, and KEC relates to capacitanceboth internal to and attached to a port according to this disclosure.Both inductance and capacitance may vary, and capacitance variations maybe sudden and large in “hot-swap” applications that are now common. Toaddress this situation a power conversion engine according to thisdisclosure may comprise circuitry Lcalc and Ccalc for determininginductance and capacitance respectively to provide “L” and “C” sub-termsfor energy terms, as is shown in this figure. It is usually practical touse the chopping waveform of this converter itself as the signal sourcefor such reactance determinations.

According to this disclosure, represented by the block ADAPT of FIG. 3,it is often adequate and even advantageous to adapt to changes ofreactances using a servo-loop implicitly responsive to L-C ratio ratherexplicit calculations based on the values of L and C themselves. Suchadaptation yields excellent response to line and load transient.Response to sudden reactance changes is slightly slower, but fast enoughto avoid transient component overloads, and usually fast enough to avoiddata loss in most digital systems.

The mathematics of energy balancing according to this disclosure andpractical means for embodying the principles thereof are fully explainedin U.S. patent application Ser. Nos. 11/593,698 and 11/593,702 filedNov. 6, 2006, which are fully incorporated herein by reference. Examplesof circuits appropriate for the KEL, KEld, KEC, Lcalc, and Ccalc,portions of calculator 150 are depicted and fully explained in theseapplications as well.

FIGS. 4A-4J depict different modes of a switching arrangement for thebidirectional and bipolar transfer of energy between non-isolated portsusing an inductor L and six switches S1 through S6. FIGS. 4A-4J showone-half of a matrix, the other half of which is identical to theportion shown, save that all polarities and all arrows depicting currentflow are to be reversed.

For all the functions shown in FIGS. 4A-4J to be performed in allpolarities, with all the modal flexibility shown in FIGS. 5A and 5B, theswitches shown must all be present and bipolar in blocking ability. Toperform subsets of the functions shown, some switches may be omitted,replaced by unidirectional switches, or replaced by diodes. Manywell-known switching power supplies and amplifiers are represented byportions of the matrix of FIGS. 4A-4J, but none embody a sufficientportion of this matrix to perform bidirectionally bipolar energytransfers.

The suffix “R” in a mode name means energy is flowing in reverse oftypical path (either from PB into inductor or from inductor out to PA).The suffix “Z” in a mode name means that the mode is used when thecurrent in the inductor is crossing zero. The suffix “E” in a mode namemeans that the mode is a reverse energizing mode. For any mode (forexample IIA, IIAR and IIAZ or IVA), whether it is the forward mode,reverse mode, reverse energizing mode or zero crossing mode, theswitches are in the same position and current is moving the same waythrough the inductor, the only difference being the test used to end themode. Certain modes may have the same switch configuration as othermodes (for example Modes IIC and IIIC) but the current is flowing indifferent directions through the inductor.

FIG. 4A, Mode IA depicts a buck converter being energized. In this modeenergy moves between ports as well as into the inductor.

FIG. 4D, Mode IIA depicts a converter transferring energy. Mode IIAperforms non-inverting forward transfers regardless of the cause ofinductive energy. The combination of Modes IA and IIA constitutes awell-known buck converter. Mode IIA is used when the PA voltage iscopolar with inductor current.

FIG. 4I, Mode IIICR depicts a converter transferring energy. Mode IIICRperforms non-inverting forward transfers regardless of the cause ofinductive energy, just as does Mode IIA. Mode IIICR is used when the PAvoltage is not copolar with inductor current.

FIG. 4D, Mode IIAR depicts a converter energizing its inductor byremoving energy from a port to lower the voltage at that port. In thismode the converter is inverting. The switch configuration of Mode IIARis identical to Mode IIA, but the test for ending the mode is based uponthe output voltage becoming less than the input voltage. Mode IIAR is anenergize mode, but is responsive to transfer strobes.

FIG. 4D, Mode IIAZ depicts a converter energizing its inductor byremoving energy from a port to lower the voltage at that port. To beginthis mode, the converter is non-inverting. However during this mode theoutput voltage will likely pass to, or even through zero. The switchconfiguration of Mode IIAZ is identical to Modes IIA and IIAR, but thetest for ending this mode is the depletion of inductive current. ModeIIAZ uses quasi-resonant behavior of the inductor acting with an outputcapacitor to pass the energy of undesired-polarity output voltage intothe inductor. When the voltage reaches zero, current in the inductorcharges the output capacitor to reverse its polarity. Mode IIAZ is anenergize mode, but is responsive to transfer strobes.

FIG. 4B, Mode IB depicts a flyback or inverting converter beingenergized. Mode IB is used for both non-inverting and invertingenergization. In this mode energy moves only into the inductor. Thecombination of Modes IB and IIA constitutes a well-known non-additive,or “bridging”, flyback converter. Modes IB and IIC form a well-knowninverting converter.

It should be noted at this point that the buck converter withouttransformer action is incapable of boosting voltage. The prior artflyback converter without transformer action can produce voltages withinthe range of the buck converter, but incurs the difficulty that very lowvoltages require very long times to transfer all energy from theinductor. With limited time available for transfer, this use tendstoward continuous conduction, which in prior art has been a relativelyvolatile mode of operation for a flyback converter. Prior art flybackconverters tending toward continuous conduction may draw large currentswhen a capacitive load must be driven away from zero volts, whichproblem engenders the need for many well-known “soft-start” circuits.The well-known “buck-boost” converter, implementing buck modes for lowervoltages and flyback modes for higher voltages, is a good choice for thesmooth unipolar, unidirectional charging of capacitive loads.

FIG. 4C, new Mode IDR transfers inductive energy into PA to return sameto the energy source when the current in the inductor and the voltage atPA are copolar. This allows for the efficient draining of energy fromthe inductor when the energy is not needed by the load at PB. Returnedenergy to the energy source at PA may result in more efficient use ofoverall energy. Where the PA energy source is a battery the returnedenergy may recharge the battery. Where the PA energy source is a powersupply the energy may be stored in the supply output capacitor orotherwise used by the power supply to reduce the overall energyrequirements of the system.

FIG. 4J, Mode IVA is just like Mode IDR, but is used when the current inthe inductor and the voltage at PA are not copolar.

FIG. 4J, Mode IVAE is an auxiliary mode that works just like Mode IVA,but is used when the next mode will be intolerant of non copolarcurrent. This occurs mainly in straddling situations when converterinput voltage has changed polarity but converter output polarity hasnot. Mode IVAE is ended when inductor current becomes insignificant. Itshould be noted that Mode IVAE is a reverse energizing mode. Mode IVAEis provided to prevent possible endless loops in the flowchart should awrong-way current be encountered for a short time when crossing zero,but this infrequent condition is difficult to invoke, and even thenwould last for but a short time. Though a mode is difficult to invoke,its absence could incur a converter malfunction.

FIG. 4E, Mode IIC performs inverting forward transfers regardless of thecause of inductive energy when the current in the inductor and thevoltage at PA are copolar.

FIG. 4H, Mode IIIBR is like Mode IIC, and performs inverting forwardtransfers regardless of the cause of inductive energy when the currentin the inductor and the voltage at PA are not copolar.

FIG. 4E, Mode IICR depicts a converter energizing its inductor byremoving energy from a port to lower the voltage at that port. In thismode the converter is non-inverting. The switch configuration of ModeIICR is identical to Mode IIC, but the test for ending the mode is likethat of IIAR, to which this mode is analogous. Mode IICR is an energizemode, but is responsive to transfer strobes.

FIG. 4E, Mode IICZ depicts a converter energizing its inductor byremoving energy from a port to lower the voltage at that port. To beginthis mode, the converter is inverting. However during this mode theoutput voltage will likely pass to, or even through zero. The switchconfiguration of Mode IICZ is identical to Modes IIC and IICR, but thetest for ending this mode is identical to that of Mode IIAZ, to whichthis mode is analogous. Mode IICZ is an energize mode, but is responsiveto transfer strobes.

FIG. 4F, Mode IID performs inverting forward transfers regardless of thecause of inductive energy when the PB voltage, VO, is small. By placingPB in series with PA it avoids unduly long times that might otherwise berequired to de-energize the inductor at a low voltage. Mode IID is usedwhen the PA voltage is copolar with inductor current.

FIG. 4G, Mode IIIAR performs inverting forward transfers regardless ofthe cause of inductive energy when the PB voltage, VO, is small, just asdoes Mode IID. Mode IIIAR is used when the PA voltage is not copolarwith inductor current.

The modes of FIGS. 4A-4J, together with those of its correspondingpolarity and current-reversed matrix-half, comprise a bipolar,bidirectional, power-conversion engine capable even of being used as anamplifier with regenerative load-energy recovery. Subsets and supersetsof this matrix may be used to form amplifiers, motor-controllers,frequency-converters, and many types of power-supplies, somepresently-known and some novel. Prior art bidirectional converters andbipolar converters use subsets of FIGS. 4A-4J, but are not bothbidirectionally bipolar in operation. It should be noted that thoughFIGS. 4A-4J show six switches, S4 is used only in new modes that improveperformance. Either S4 or S1, and modes using same, may be omitted topractice bipolar, bidirectional power conversion according to thisdisclosure. It should further be noted that some of the switches ofFIGS. 4A-4J may be replaced by diodes to embody subsets of its modematrix. Some of these sub-sets correspond to well-known power convertersand other subsets are new.

FIGS. 5A-5D show different portions of a flowchart depicting thesequencing of the hardware functions shown in FIGS. 4A-4J necessary toimplement a bipolar, bidirectional, buck-boost, energy-balancing,synchronous power-conversion amplifier using an inductor. Power suppliesand amplifiers that may be unipolar, unidirectional, buck or boost,non-energy balancing, or asynchronous, including well-known prior artapparatus, are partial implementations of the hardware matrix of FIGS.4A-4J and correspondingly partial implementation of the flowchart ofFIGS. 5A-5D.

An arbitrary rule convenient for operation of the apparatus of FIGS.5A-5D is that whenever a particular phase of operation has beencompleted, a series of tests is initiated to determine whether energyexists in the inductor L of FIGS. 4A-4J and, if there is, whatsubsequent function best accomplishes a desired power-conversionfunction and, if there is not, whether it is desirable to energize saidinductor. If an energy balancing conversion is performed, suchenergization will be responsive to an energy balance term that indicateswhether such energization has become adequate for that mode or asubsequent transfer mode.

Operation may arbitrarily be traced from the point marked LOGIC 1 at thetop of FIG. 5A. Though illustration of the functions of FIGS. 5A-5D as aflowchart eases understanding, and is naturally adapted to a digitalprocessor, it should be understood that FIGS. 5A-5D may also represent aregistered memory containing a table of states, with only one of thefifteen modes shown allowed be active at any time. Implementation ofFIGS. 5A-5D by such means as a processor requires that care be taken toavoid intermediate illegal states or combinations of states that mightcause power converter runaway or a loop of operations lacking an exit.In FIGS. 5A-5D, the modes labeled indicate the correspondingly labeledelements of the matrices of FIGS. 4A-4J.

A path to logic 1 lost shown on FIG. 5B begins with test 1 to determinewhether or not significant VO is present. With no VO, test 2 determineswhether significant VI is present. The absence of both indicates thatnothing is to be done to VO, but since L may contain energy, test 23 isneeded. If L is exhausted, nothing need be done. However, if L hasenergy it should be returned to VP, the energy source attached to PA. Ifthe current in L is copolar with VP, test 24 selects Mode IDR to returnthat energy, otherwise Mode IVA is used for the same purpose. As apractical matter the test see whether there is any VO, VI or inductorcurrent compares theses values to a small threshold value, and if thevalue is less than the threshold, it is approximated as zero.

Turning again to test 2, if VI exists and VO does not, energy will beneeded to increase VO. If VI and VP are non copolar, some invertingmeans of getting energy to VO is needed, and test 3 is followed by test4 to determine whether L has energy to contribute. If it does, test 5determines the copolarity of that current with VP, and if non-copolartest 6 determines whether VP is much larger than VO. Since we justfollowed a no-VO path, this will be true and Mode IIIAR will beselected. Had we arrived here by another path with a larger VO, ModeIIIBR would have been selected. Had test 5 determined copolarity, eitherMode IIC would have replaced Mode IIIBR or Mode IID would have replacedMode IIIAR.

Turning again to test 4, if L had no current, Mode IB would be used toenergize it to prepare for a likely future inverting forward transfer byMode IIIBR, IIIAR, IIC, or IID. When the current in the inductor risesabove the threshold current the result of test 4 switches to “no” andreturns to test 5 as described above.

Turning again to test 3, had VI been copolar with VP, test 11 woulddetermine whether L has energy. If it does, test 13 determines whetherits current is copolar with VP. If it is not copolar, Mode IIICR will beused to move that energy usefully to PB, but if it is copolar, or iftest 11 determines it to be zero, Mode IA will vigorously slew VO. Sincea no-VO path lead here, there is no danger that VO exceeds VP.

Returning to test 1, if VO exists test 14 determines whether VI exists.If there is VO but no VI it will be necessary to remove energy from PBand return it to P. To do this, test 20 determines whether L hascurrent. If it does not, test 22 will select either Mode IICZ or IIAZ,depending on the copolarity of VO with VP to move energy from PB to L,which will continue until IL has grown and then shrunk to zero, and VOhas been reversed in polarity, or until VO has attained equality withVI.

Returning to test 20, if L has current that is copolar with VP,subsequent performance of Mode IICZ or IIAZ is selected by test 22 aswith no current, but will be accomplished more quickly. However noncopolar IL and VP would malfunction in Modes IICZ and IIAZ, so in thisevent test 21 selects Mode IVAE to return this ill-polarized energy toPA.

Returning to test 14, there may exist a VI, which may or may not becopolar with VO. If test 15 determines non-copolarity, a straddlingcondition exists which, proceeding to test 20, is handled like a no-VIcase. However, whereas a no-VI case would require VO slewing only tozero, the straddling case exploits the quasi-resonant polarity reversingabilities of Mode IICZ and IIAZ smoothly to slew VO to the other side ofzero. It should be noted that L and any C at PB, when switch-connected,form a lumped element transmission line with a finite cutoff frequencyunable to pass a fast-slewing waveform. The quasi-resonant method yieldssmooth slewing with the fastest practical slew rate.

Returning to test 15, if straddling is absent, test 16 determineswhether energy must be moved to or from PB. If VO is less than VI,energy must be added to PB, which leads to test 8. Test 8 determineswhether inversion is needed, and if so, the same path through test 4occurs as when no VO was present. If, however, no inversion is needed,test 8 leads to test 9 to see whether VO is greater than VP. If VO isgreater, a flyback mode must be used, leading to test 12. Test 12determines whether L has energy to contribute to PB, and if it does notL must be energized by Mode IB. If L has energy, test 12 leads to test10 to determine copolarity of IL with VP. Copolarity leads to deliveryof the energy of L to PB through Mode IIA, and non-copolarity leads toanalogous delivery through Mode IIICR.

Returning to test 9, a small VO leads to test 11 to ascertain IL, in theabsence of which it is appropriate to invoke Mode IA. However, if ILflows, test 13 determines whether IL is copolar with VP. If it is notcopolar, Mode IIICR is used to move the energy of L to PB. If IL iscopolar, Mode IA slews VO vigorously.

FIGS. 5A-5D show two modes of forward energization, Mode IB, a flybackmode, and Mode IA, a buck mode. Since switch on-times in powerconverters are related to volt-time product, low voltages incur longtimes to move inductive energy. Thus the flyback mode is preferable tothe buck mode when a generator port voltage exceeds a dissipator portvoltage and vice-versa. With a simple, two-terminal inductor converterno way exists to make the same inductor simultaneously a current sink toa dissipator port and a current source to a generator port of oppositepolarity. Therefore the buck mode is limited to energy movements wherethe dissipator and generator ports are copolar. Accordingly, all FWDnon-buck energizations share energizing Mode IB. It should be noted thatbuck Mode IA possesses simultaneous EN and TR energy moving ability. (ENmeaning energy is transferred from PA to inductor and TR meaning energyis transferred from inductor to PB.) It may be seen and should be notedthat though Mode IA works smoothly, since it cannot provide to aninductor inverting modes its use results in asymmetrical plus and minusslews below the absolute value of VP. Therefore Mode IA should beomitted and Mode IB and an appropriate transfer mode substituted for anyamplifier applications where waveform symmetry is paramount.

In a synchronous converter, the tests above the row of mode registers ofFIGS. 5A and 5B are continually being performed, so the data on theregister inputs represents the best mode for the next upcoming clock ortrigger. The timing generator logic comprises priority circuitry toallow energize Modes IA, IB, IVAE to be selected only at the time of asynchronous clock. The strobes for each mode register set may comprisean update strobe to latch in new data, and a reset strobe to de-select aregister set. At synchronous clock time, any mode may be selected. Atthe time of a first asynchronous trigger, modes other than IA, IB, andIVAE may be selected. At the time of a second asynchronous trigger, onlyreverse transfer modes IVA and IDR may be selected and all other modesare reset.

With an energy demand and a lack of inductive energy, the appropriatetask is to energize the inductor, but with the inductor energized, itsenergy needs to be moved to a port using a forward transfer mode, IIA,ITC, IID, IIIBR, IIIAR, or IIICR, any of which may commence either atsynchronous clock or at first asynchronous trigger. If excess inductiveenergy exists, the forward transfer mode may be terminated and a secondasynchronous trigger generated. Also, a forward transfer mode may beskipped. At a synchronous clock, at a first asynchronous trigger, or ata second asynchronous trigger, a reverse transfer mode may commence toreturn excess inductive energy to the power source attached to VP. Areverse transfer may end in one of two ways. One way is exhaustion ofinductive energy, after which the inductor may rest de-energized until anext cycle begins. A second way is that the time allotted for this modehas ended. It should be noted that a single cycle of a synchronousconverter of this disclosure may comprise many intervals in whichdifferent modes are selected to facilitate the most timely and accuratedisposition of energy in response to port demands and conditions. (Threedifferent modes are not meant to be a limitation and the use of moremodes is within the scope of the disclosure.) An aspect of thisdisclosure shown in this figure provides for three selected modesfollowed by a default idle mode, the latter sometimes being ade-energized occurrence of another mode such as Mode IDR. Such a defaultidle mode actually corresponds to the idle or null mode of prior artconverters. Thus, during a conversion cycle, this disclosure, unlikeprior art converters, may comprise four modes, three of which may moveenergy. The first two, an energizing mode and a forward transfer modeare usual in the prior art. The inclusion of a reverse transfer modewithin a converter cycle of this disclosure contributes to its excellentefficiency, transient response, and low input and output ripple. At thispoint it is necessary to emphasize a fundamental difference between thisdisclosure and prior art. Prior art power converters usually follow muchsimpler rules in transferring energy. In most converters a transfer modefollows a energizing mode and vice-versa. According to this disclosure,whenever a mode ends a new decision is made about how best to moveenergy within the converter.

The power conversion processes of FIGS. 5A-5D may be implemented with orwithout energy balancing. If energy balancing is not used, the testspertaining thereto are simply omitted. Omission of energy balancingrequires a feedback loop based on VO filtration rather than prediction,and incurs a pole in the feedback loop, which tends toward impreciseinductor energization in any given cycle, and often engenders inmulti-cycle correction and consequent sub-harmonic ripple on VO.

Just as FIGS. 5A and 5B illustrate the logical decisions requiredappropriately to initiate modes, FIGS. 5C and 5D illustrate both thedecisions needed to terminate modes and the timing of such mode endingsneeded to facilitate orderly progression to a next appropriate mode.Modes 3BR, 3AR, 2C, 2D, 2A, and 3CR are all complete when the current inthe inductor is exhausted. Therefore, test 25 initiates a secondasynchronous trigger after exhaustion of inductor current in any ofthose modes. A second asynchronous trigger evokes a Reverse TRansferSTroBe, but that strobe can initiate Mode IDR or 4A only if inductorcurrent exists, so these modes are not invoked. If inductor current isnot exhausted, test 26 monitors whether VI exceeds VO. If it does not,VO is unnecessarily large, so the inductor energy is not needed at theoutput port, and will be returned to VP when the second asynchronoustrigger and the RTR STB invoke either Mode IDR or IVA. However, if VIexceeds VO the original mode will continue.

If the power converter includes energy prediction, Mode IB will beterminated upon attainment of forward balance by test 30 to generate afirst asynchronous trigger which can make either an Forward TRansferSTroBe or an RTR STB, initiating some kind of transfer form theinductor. If prediction is absent or balance is not attained, test 28 ismade for Turn-Around which also can result in a first async strobe whichwill end the energizing Mode IB and invoke some sort of transfer mode.Likewise if TA is not reached, but a current limit is reached, test 29aborts Mode IB.

Mode IA is a buck mode and may use balance test 31. If balance occurs,the path is the same as just described for Mode IB balance. However ifbalance is not reached, or is absent, test 27 monitors whether VO hasgotten too large. If it has, Mode IA is ended as previously described.If it has not, tests 28 and 29 continue to monitor their variables asdescribed above and stand ready to end Mode IA.

Mode IVAE occurs infrequently and is provided to prevent the converterfrom getting caught in an endless loop. Mode IVAE is initiated only atthe start of a chopping cycle and, through test 34, results in a firstor second asynchronous trigger depending respectively on whether or notinductor current has become insignificant.

Test 32 generates a second asynchronous trigger whenever Mode IICR orIIAR reduce VO to less than VI, invoking Modes IDR or IVA as describedabove.

Modes IICZ and IIAZ are terminated by test 33 when an undesired VO hasbeen substantially eliminated.

Both a synchronous and an asynchronous timing generator are shown inFIGS. 5C and 5D, only one of which would be used at one time in anygiven converter. Both function similarly accepting a beginning clock andtwo asynchronous events. In the synchronous case, the beginning is asynchronous clock, but in the asynchronous case the beginning is merelythe delayed end of the last chopping cycle. The energize Modes IB, IA,and IVAE may occur only at clock time. If they do not occur an FTR modemay be initiated instead. If an FTR mode does not occur, an RTR mode mayoccur. In like manner, either an FTR or RTR mode may occur at a firstasynchronous trigger, but not an energize mode. A second asynchronoustrigger may evoke only an RTR mode. This priority of action is enforcedby circuitry of the timing generator.

The action of this timing generator working with the modes of thisdisclosure produces novel behavior. Most prior art power converterscommonly execute energize modes and forward transfer modes. Many alsohave a default null or “do-nothing” mode exemplified by the state of thediscontinuous flyback converter after the inductive energy has beenexhausted.

It should be noted that in Mode IB, VO will not respond to the energybeing imparted to the inductor until another mode is selected. If aprior art feedback loop be attached to VO, that feedback loop will beopen until Mode IB is done. According to prior art, energy will bedispensed based on the history of previous cycles held in an outputfilter, and a pole will occur in the feedback loop. However, if theenergy-balance generator of FIG. 10 described below be employedaccording to this disclosure, predictions made thereby willasynchronously terminate Mode IB when the correct amount of energy hasbeen imparted to the inductor. Then a subsequent transfer will result inVO arriving at substantially the correct voltage desired with no excessor lack of energy, obviating the need to correct VO in subsequentcycles.

More complex and less complex forms of the disclosure of FIGS. 4A-4J and5A-5D are practical. Many incomplete forms are novel and others exist inthe prior art. For example, the well known buck converter is a sub-setof FIGS. 4A-4J and 5A-5D, using only Modes IA and IIA. Unlike prior artconverters, FIGS. 4A-4J and 5A-5D depict a new power conversionapparatus and method respectively, that implement bipolar, bidirectionalenergy transfers, making possible not only power supplies, but alsoswitched mode power amplifiers, and other useful devices. Moreover, thebidirectional energy-moving ability of this disclosure facilitatesreturn of excess energy to a power source, facilitating power convertersof higher overall efficiency than prior art means. Not only that, butextension of the principles of FIGS. 5A-5D, for example, switching sensemeans from PB to PA of FIGS. 4A-4J to exchange active and passive portsenables not only bidirectional energy transfer but also bidirectionalregulation. Even more capability can be added by using the sameprinciples with more than two energy-moving ports.

While it is practical to embody this disclosure to perform in awell-known continuous current mode (CCM), as is described below in FIG.14, adding complexity of that function to the already complex functionof FIGS. 5A-5D might prove unwieldy, so this discussion of FIGS. 5A-5Dis limited to the well-known discontinuous current mode (DCM).

It should be noted that FIGS. 5A-5D represent but one of many practicalalgorithms for operating the switches of FIGS. 4A-4J. Other algorithmsare practical, for example a CCM converter in which the inductor oftencarries a reservoir of excess energy can improve transient response withbut a slight efficiency loss.

FIGS. 6A-6C depict different modes of a bidirectional switch 300suitable for use in the present disclosure in which two MOS switches 301and 302 are disposed so that their body diodes block in both directions;NMOS or PMOS switches could alternatively be used.

It should be noted that FIGS. 6A-6C depict a convenient, but notnecessary, way to embody switches according to this disclosure, BJTs,IGFETs, thyristors, magnetic amplifiers also being practical or anyother type of power switch, direct coupled or isolated, whether nowknown or hereinafter invented, may be used. In certain embodiments,where all modes are not necessary, switches may be eliminated witheither an open or closed circuit, or switches may be replaced by adiode.

FIGS. 6A-6C are divided into three sub-figures, FIGS. 6A, 6B, and 6C.FIG. 6A also depicts circuitry responsive to the current IL in theinductor L. Referring first to FIG. 6A, there is shown current comparingcircuitry 200 including a current, or other inductive energy, sensor 201which generates a signal VIL that is used within 200 and in FIG. 10below. Comparator 202 compares VIL with a threshold source 203 and, ifIL is insignificant generates the signal IL<th used in the tests ofFIGS. 5A-5D. The signal IL<th flows into switch 300 where switch logic303 uses it to control transfer modes. Switch logic 303 is fitted withtwo control inputs, A and B for individually controlling the FETs 301and 302. For transfer modes these inputs A and B are individuallycontrolled, but for energize modes both are usually simultaneouslyexerted.

Referring to FIG. 6A and to FIG. 7, switching is preceded by an off timeduring which no current would flow even if FET 301 of FIG. 6A were on.Most likely this time is an energizing mode. It is practical to turn onFET 301 during this time to avoid having no current path for theinductor should the present mode cease.

When conduction from another mode ceases, inductor current continues,causing VL to swing positive, initiating the conditions shown in FIG. 6Acorresponding to TIME a of FIG. 7. Current IL flows in the body diode ofFET 302, during which time VL rides about one diode-drop above loadvoltage Vld. This current is sensed by the current sensor 201 andcompared with a current threshold 203.

When current IL exceeds said threshold, switch logic 303 initiates thecondition shown in FIG. 6B, corresponding to TIME b of FIG. 7 by turningon FET 302 which now acts as a synchronous rectifier, reducing diodelosses.

When inductive energy is nearly exhausted and current IL drops belowsaid threshold 203, switch logic 303 initiates conditions depicted inFIG. 6C, corresponding to TIME c, by turning off FET 302. Conduction nowresumes in the body diode of FET 302 until residual energy is exhaustedor until another mode turns off FET 302. TIME c of FIG. 7 is the stateof having attained the end of the condition “until Ith” and would resultin a yes Y decision in any |IL|<Ith test of FIGS. 5A-5D.

The temporal separation of the three conditions of FIGS. 6A-6C and 7prevents undesirable simultaneous conduction, or “shoot-through”, whenswitching between the modes of FIGS. 4A-4J. With the switch timing ofFIG. 7 used for transfers as just described, there is but little time inwhich a circuit around the inductor is incomplete, minimizing energyloss in snubbers.

FIG. 7 depicts as waveforms the switching action of FIGS. 6A-6C as itpasses through the three conditions, as just described.

FIG. 8 depicts a practical switch arrangement for the switches of FIGS.1 and 4A-4J. Pairs of FETs 3011-3021 through 3016-3026, corresponding toS1-S6 of FIG. 1 and 301-302 of FIGS. 6A-6C, and having body diodes (notshown), embody the needed bi-directional switch function betweeninductor L and ports PA and PB, as is also shown in FIGS. 1 and 4A-4J.The need for a pair of FETs for each switch is a consequence of thecommon practice of reducing terminals by tying the body terminals ofmost large FETs to their sources, causing them to blockunidirectionally. Should this practice change to produce bidirectionallyblocking FETs, single FETs rather than pairs of FETs would suffice topractice this disclosure.

Since practical FETS may be capacitive, commercially available FETdrivers shown included within IS1-IS6 a and b may be used to drive them.Voltage sources VA through VG are used to power such drivers through thenode pairs A-A+, B-B+, C-C+, D-D+, and G-G+.

Though AC coupling may be used to drive FETs, such coupling oftenmalfunctions at extremes of switch duty-cycle. Isolated D-C coupling isusually most reliable. The isolation also shown included within IS1-IS6a and b is preferably accomplished with Gross Magneto Resistive (GMR)isolators capable of DC operation. Optical isolators may be used wherespeed is not important. Transformer-coupled data isolators such as thoseof the ADUM series from Analog Devices may be used, but at the risk ofundesired bistable states. Though FETs are commonly used in powerconverters, BJTs, IGFETs, thyristors, or even magnetic amplifiers may beused as switches to practice this disclosure. Referring to IL, RIL andAIL of FIG. 8 for providing a signal VIL representing inductive currentin L, in some cases a simple amplifier will suffice, but it must possesssufficient GBW (gain bandwidth), input range, and DC accuracy for thetask. Commonly available high-side current monitors usually haveinsufficient bandwidth for high speed converters. A current transformerproperly summed with a current monitor can provide flat response from DCto a high frequency. A current transformer alone provides only ACcoupling, which can be made to work in limited cases, but incurs therisk of undetected high currents.

FIGS. 9A and 9B depict the converter-based generation of a sine wavefollowing a control port voltage, VI, starting positive from zero atzero degrees. These graphs were generated by a SPICE simulatorsimulating a circuit embodying FIGS. 4A-4J plus logic embodying thefunctions depicted in FIGS. 5A-5D. FIG. 9A shows operation with apositive PA voltage, Vp while FIG. 9B shows operation with a negative PAvoltage. The current ebbing and flowing in PA, Ip, is shown at the topof FIGS. 9A and 9B. In the middle are shown a control port voltageinput, VI and, VO the sine wave voltage produced at PB by the powerconverter of this disclosure being simulated.

Just below these analog waveforms is a timing diagram produced by theSPICE simulator as it was simulating this converter. The designations tothe left of the timing diagram, 1A, 1B, etc., show which of the modesdescribed above is active at any given time and causing and respondingto the waveforms above the timing diagram. (The Arabic numbers of FIGS.9A and 9B generated by SPICE correspond to the Roman numerals of FIGS.4A-4J and the description below; thus Mode 1 and Mode I are the same.)

Starting with FIG. 9A, Mode IA, the buck-energize mode, first drives upPB voltage, VO, across a energy-storing load connected thereto. Sincethis is a synchronous embodiment, an arbitrary rule allows Modes 1A, 1Bto be initiated only by a clock pulse, in this case supplied at 20 uSintervals (not shown). Since Mode IA vigorously slews VO, the latteroften slightly overshoots VI as it touches same, indicating excessinductive energy, and causing an occurrence of Mode IDR to return energyto PA. As VO approaches VP, here +5V, the vigor of slewing decreasescausing VO not to overshoot VI, which first happens at 80 uS, at whichtime a Mode IIA moves inductive energy to PB. Upon exhaustion of theinductive energy, Mode IDR is invoked, which can either to return energyto P or, in this case simply load the exhausted inductor, L.

At the 100 uS clock (not shown), VO is now about equal to VP, or about5V, so Mode IA would no longer slew VO. Therefore Mode IB now begins,and lasts one 20 uS clock cycle, energizing L. At the 120 uS clock, ModeIIA moves inductive energy to PB. When VO is nearly to VI at about 126uS, an asynchronous sequencer decision invokes Mode IDR to return excessenergy to PA. The current drawn from VP (pointing down, can be seen foreach occurrence of Mode IB. This sequence is repeated at 140, 180, and220 uS.

Since VO is now very close to VI, invocations of Mode IIA becomesshorter while invocations of Mode IDR increase in length.

At 250 uS the 90 degree sine wave peak is reached, and new modes begin.At 260 uS, VO is above VI, so Mode IICR draws energy from PB into Luntil VO descends to VI, at which time another sequencer decision againinvokes Mode IDR. Energy taken from PB now resides in the inductor, L,and Mode IDR returns that energy to PA, making an upward-pointing spikeof VP current. This spike ends either when inductive energy is exhaustedor at the next clock. This alternation of Modes IICR and IDR continuesfor almost 90 degrees of the sine wave, with additional alternationscommencing at every 20 uS clock until 500 us. Additional spikes ofreturned current may be seen between 90 and 180 degrees of the sinewave.

At about 500 us VO is positive and lagging VI which has already gonenegative. We have encountered the first occurrence of straddling zero.At this time Mode IICZ is invoked, dumping now-excess energy from PBinto L. L, in turn, quasi-resonantly recharges a capacitor at PB,returning much of its energy, but in the opposite polarity. One can see,just after 500 uS, VO passing through zero to arrive several hundred mVnegative under the influence of Mode IICZ.

Just before 520 uS, a tiny spike of Mode IID gives evidence of a littleexcess energy in L. The reason that Mode IID occurs rather than IIC isthat VO is much less than VP.

Less than perfect timing causes the opportunity at 520 uS for Mode IB tobe missed, but at 540 uS Mode IB occurs, energizing L. At 560 us anadditional Mode IID pulse begins and is asynchronously terminated toinvoke Mode IDR when L becomes exhausted. At 580 uS an new Mode IBstarts, but at 600 uS when its new energy in L must be moved, Mode IICis invoked, because VO is no longer tiny compared to VP. Until 700 uSeach Mode IIC is followed by a Mode IDR, not from VO reaching VI, butfrom exhaustion of L1. After 700 uS there is a Mode IDR accompanied by areturned energy spike of Ip. Of course, each Mode IB was accompanied bya spike of energy provided by PA. At 750 us, 270 degrees of the sinewave is reached and both VI and VO are close together and quite flat. Lis full of excess energy that gets returned by Mode IDR in a relativelylarge spike at 760 uS.

At 780 uS Mode IIAR first begins. This mode, like Mode IICR moves energyfrom PB to L, until VO rises to meet VI, at which time a sequencerdecision again evokes Mode IDR to return energy, evidenced by an Ipspike, to PA. Alternations of Modes 2AR and IDR continue, with returnenergy spikes, commencing with every 20 uS clock pulse until 1 mS.

At about 1 mS VO is negative and lagging VI which has already gonepositive. We have encountered the second occurrence of straddling zero.At this time Mode IIAZ is invoked, dumping now-excess energy from PBinto L. L, in turn, quasi-resonantly recharges a capacitor at PB,returning much of its energy, but in the opposite polarity. One can see,just after 1 mS, VO passing through zero to arrive several hundred mVpositive under the influence of Mode IIAZ.

Next Mode IA recurs to slew VO up the rise of a new sine wave cycle.

Referring to FIG. 9B, a similar sine wave generation occurs, but with VPnow at −5V.

The modes used are similar and have similar functions, but with somerole-reversals. Mode IA being a buck mode cannot be used for inversion,so it now occurs when both VP and VI are negative. Some energy-returningIDR modes are now replaced by VIA modes because of polarity reversal.The negative slews of FIG. 9B correspond to the positive slews of FIG.9B. Energy use now points upwards on Ip and energy return pointsdownward.

A cursory glance with understanding of the larger peaks of Ip shows theadvantage that could be gained by including energy-balancing into thisconverter. The large spikes are associated with Mode IB. Lackingtermination by an energy-balance, Mode IB continues until cycle's endunless a current limit is reached. Then during the next clock cycle,energy is dispensed and any excess returned to PA.

Were energy balancing in place, ModeIB would asynchronously terminateupon attainment of sufficient energy in L, allowing energy dispensing tocommence, and perhaps even be completed within the same cycle. Theenergy return spikes interspersed with Mode IB energizing spikes woulddisappear, and the energizing spikes would be smaller. Both voltageripple on VO and current ripple in VP would be reduced. It should benoted that any energizing mode of FIGS. 4A-4J may be followed by anytransfer mode. What is important is not how energy got into the inductorbut which transfer mode best moves that energy to accomplish a desiredoutput with utmost efficiency.

FIG. 10 shows an energy-balancing calculator that may be used as part ofthis disclosure to improve performance. This balancing apparatus may beomitted from embodiments of this disclosure wherein energy-balancing isnot practiced. It may also be implemented by implicit, adaptive means asshown in FIG. 12 rather than by the explicit means shown here forclarity. It should be understood that means for determining values ofcapacitance and of inductance, and of indirectly determining loadcurrent may be added to this calculator as taught in related applicationSer. No. 11/593,702, incorporated by reference herein. It should benoted that though mathematically correct implementation of the energybalancing calculation taught in this figure produces the excellentresults, crude approximations of the predictions thereof still offersuperior results to non-predictive prior art control. Thus it ispossible to practice this disclosure by using simplified predictiveterms, for example by neglecting to square quantities in the KEL and KECterms, to produce results superior to the prior art.

There are three main parts in the calculator of this figure, the firstof which is the KEC, or capacitive kinetic energy, term.

VI, suitably conditioned for polarity, is squared in multiplier BM1; VOis likewise conditioned and squared in multiplier BM2, and then invertedby inverter BI1. Summing of the first square and the inverse of thesecond is performed by BS1 to yield a difference of squares, which isrectified by BRECT1 and then multiplied in BM3 by 0.5 times the value oftotal capacitance CTOT at the converter output. This process yields theenergy that will be requires to move VO to a desired voltage. The signof this move is addressed using polarity signals.

The second, inductive kinetic energy, term KEL is more complex. Itssimplest part is the instantaneous inductive energy, KELt, obtained asfollows. BM6 squares IL (practically represented by VIL in FIG. 8) andmultiplies the same by 0.5 L, the inductor value/2, providinginstantaneous kinetic energy in accordance with the well known equationKE=L*I^2/2. Since it takes time and voltage to extract energy from aninductor, there is another inductive energy term KELd (not to beconfused with KEload) that represents the de-energized energy of theinductor at the end of a mode. If the inductor had time and voltagefully to de-energize, KELd is zero, but if not KELd needs to bedetermined by computing ILd, the de-energized inductive current. Toobtain ILd, BM4 first multiples VO by dT, the time remaining in themode. The term dT may be a descending sawtooth wave representingremaining cycle time. BD1 then divides the resulting product by L. Thisaction implements the well-known equation dI=E*dT/L. The result, dIL isinverted by BI2 to obtain −dIL, which is then summed by BS2 with VIL toyield ILd. which is rectified by BRECT2. The resultant signal is squaredand multiplied by 0.5 L to yield de-energized energy KELd which,inverted by BI3 is summed by BS3 with KELt to yield predictedextractable energy, KEL.

The third, load energy, term predicts how much energy will be needed tocause VO to be correct at a desired time. In the buck converter theprediction is preferably to cycle end, and the desired voltage isusually buried in the VO ripple. The same time may be used for theflyback converter, in which case the desired voltage is nearer thebottom of the VO ripple. Alternatively, the time, Tfb, till the end ofinductive flyback may be used, putting the desired voltage near the topof the VO ripple. BM9 produces the KELoad term representing thepredicted energy consumption of a load. To do this, the chosen time, dTor Tfb is multiplied by VO and by load current. Load current might bedirectly measured but, for best efficiency, it is usually preferable toderive Iload. If a known capacitor, CREF, be in shunt with VO and withany other capacitance, internal or external, ripple current will flow inthese capacitances in inverse proportion to their respectivecapacitances. Thus, by well known ratiometric techniques the value ofCTOT may be determined. If all or part of CTOT is external, it may beinconvenient to measure its current, but CREF, being internal, allowsits current Jr to be measured. Ir multiplied by Ctot in BM8 and dividedby Cr in BD3 yields total ripple current, Irip. Since most of the ACcurrent of IL is Irip, IL-Irip is the derived instantaneous loadcurrent. B14 inverts Irip for summation in BS5 with IL to yield aderived Iload term for BM9.

The balance generator of this figure also embodies additional functions.Cdif and Rdif generate a positive signal whilst KEL is increasing, but anegative signal when KEL decreases, which comparator BC4 processes to aturn-around signal TA2. Early in an EN mode while IL rises, KEL alsorises, but as time for inductive energy transfer decreases later in themode, KEL begins to fall despite the continued rise of IL. At that time,nothing but dissipation is gained by additional energizing, so TA2issues as an asynchronous trigger to end the EN mode. It should be notedthat this term according to this disclosure is applicable to convertersoperating in the well-known DCM mode. For either energy balancing ornon-energy balancing converters according to this disclosure, Tfb may becompared to dT by BC3 yielding another turn-around signal TA1, toaccomplish the same end.

While in the discussion until this point it has been assumed that theinductor, L, is an inductor, it may be any type of inductive reactor,including a transformer. In fact, where it is desirable to isolate oneport from another port a transformer will be required. Such designs arewell know in to those skilled in the art of power design and will not befurther described herein. However, the switch configurations and controltechniques described herein can easily be applied to such inductivereactors and the disclosure is meant to include such applications.

FIGS. 11A-11H depict switching states for the adaptive and simplifieddemonstrative examples of this disclosure described hereinbelow. Thesestates are similar to the modes of FIGS. 4A-4J, but represent theadditive form of flyback converter. (However, it should be noted thatStates IV, VI and VIII of FIGS. 11D, 11F, and 11H have no equivalents inFIGS. 4A-4J.) In as much as the following examples require no closure ofS1 of FIGS. 1 and 4A-4J, S1 has been omitted from this figure. A newswitch S7 has been added to facilitate illustration of recirculation ofinductive current as will be explained below. While the use of S7 isconvenient, it should be noted that the same result can be accomplishedin FIGS. 4A-4J by closing S3 and S5 or S2 and S4 to cause a shortcircuit across the inductor for recirculation. S7, however, isconvenient in that the losses associated with a second switch areavoided. In the examples below, fewer than all six of the switches inFIGS. 4A-4J are required.

In State I, the classic forward energize mode of additive flybackconverters, L is shown being energized through S2 and S5.

State II shows the classic flyback, or forward transfer, mode ofadditive flyback converters.

State III shows the reverse energizing of L from PB through switches S6and S3. Such energizing might be used to remove excess energy from PB.Note that inductive current flow is reversed from States I, II, and V.

State IV shows a reverse transfer of energy from L to PA. Such atransfer might be used to return to PA undesired energy from PB afterState III or State VIII. Note that inductive current flow is reversedfrom States I, II, and V.

State V shows the return of energy from the inductor, L through switchesS3 and S4. This mode might be used to return to PA excess forward energyin L.

The movement of LD and VDC in various States is not to say that the loadand power source have been moved, but only to indicate that energy isnow flowing out of the converter 100 to PA, or into converter 100 fromPB.

State VI shows recirculation of inductive current in L through S7 tomaintain a store of energy therein according to this disclosure as isexplained below. It will be clear to anyone practiced in the art that,with slight additional losses, recirculation according to thisdisclosure can also be effected by the closure of both S2 and S4, or ofS3 and S5, with all other switches being held open.

State VII shows the forward transfer of reverse energy to PB throughswitches S3 and S6.

State VIII shows the forward negative energize state for loading L withnegative energy through switches S4 and S3.

It should also be noted that the control mechanisms herein control anoutput with respect to some reference signal. The reference signal mayby the input voltage to the converter, a separate signal to adjust thecontrol, or both of the foregoing.

DEMONSTRATIVE EXAMPLES

A number of examples of embodiments of the disclosure are set forthherein. The Inventors believe that these embodiments demonstrateparticularly beneficial implementations of the techniques of thesedisclosures, but these are not meant to be limiting in any way withrespect to the scope of the disclosure.

Example 1 Amplifier Using Simple Flyback

FIG. 12 represents an amplifier using the simplest flyback converterform, comprising one switch (corresponding to S5 of FIG. 1) and onediode (corresponding to S6 of FIG. 1). This synchronous additive-flybackunipolar amplifier embodies methods of obtaining both CCM stability andadaptive self-correction according to this disclosure.

FIG. 12 comprises two major portions. On the right is an entirelyconventional additive-flyback power converter comprising inductor L,snubber capacitor C8, snubber resistor R26, inductive current sensorSEN, switch S5, S-R flip-flop BIS, over-current comparator COC,current-limit threshold VLIM, switching diode S6, filter capacitor CI,voltage divider RA and RB, over-voltage comparator COV, reset or-gateROR, and an output voltage VO, to which is connected a load LD. Save theuse of one input of gate ROR to accommodate a BAL input, the functionsof the components of the conventional portion are the same as the inprior art. What is missing from the prior-art portion of thispower-converter is the usual feedback loop attached to the input voltageVI, the output voltage VO and the aforementioned input of gate ROR toregulate output voltage VO.

Only two of the energy moving states shown in FIGS. 11A-11H, the ForwardEnergize state (I) of FIG. 11A and the Forward Transfer state (II) ofFIG. 11B, are used. A relatively large minimum load is required for suchan amplifier because it can actively slew in the positive directiononly.

A four-switch version of this amplifier has also been demonstratedshowing a symmetrical dynamic response, see FIGS. 20A and 20B. In FIG.12, S6 has been replaced by a diode to provide a simple example.

The amplifier of FIG. 12 comprises, in place of a prior-art feedbackloop, an adaptive feedback loop ADAPT according to this disclosure, thatembodies adaptive correction that can be applied to adjust for anyaggregate errors, and any additional capacitance in shunt with the loadLD. If an instantaneous input voltage, VI, is used for the adaptivecorrection, the effect is to add phase lead to an output, VO. Theability to introduce lead, stably and predictably, in reactive feedbackloops can be a useful feature in control systems, particularly thoseinvolving momentum. If lead is not desired, using matching input andoutput averaging filters allows adding adaptive correction withoutaltering phase. R3 and C2 form the input averaging filter that producesan average input voltage, Vlavg. Rvo and Cvo form a corresponding outputfilter that produces an average output voltage, VOavg. The adaptivefeedback can compare the instantaneous input to the filtered output anddraw those two voltages together. That causes phase lead. If the inputvoltage is filtered in the same manner as the output voltage, theadaptive feedback will not cause phase distortion.

This example shows a voltage gain of 1. Any reasonable gain can beapplied either digitally, or by adding a gain stage on the inputvoltage, VI, or by dividing the output, VO, before it is used in thecalculations. Any non-linear transfer function desired can be likewiseincorporated.

In this example, BIS is synchronously set every 5 uS by a clock pulsesignal S from timing generator TG, to energize L in conventional fashionby closing switch S5, admitting current from a power input, VP. Aconventional output filter capacitor is here designated as CI toindicate that it is the filter capacitance internal to this powerconverter. In this example, load LD draws a sinusoidal or pulsed currentload for testing purposes.

Sensor SEN may be a well-known current-monitor or other circuitry forascertaining inductive energy to produce a signal VIL representinginductive current. Behavioral voltage source B1 may be responsive to thevalue of L, or may be a fixed voltage representing the value of L. Aconstant voltage 0.5 is also shown. Behavioral multiplier BM6 processesthese three inputs to implement the classic inductive energy formulaKE=½L*I^2, generating an inductive-energy signal KEL.

Timing generator TG comprises two signal outputs TG1 and TG2 and twodelay elements, DL1 and DL2. TG1 is a 100 nS pulse with a 5 uS periodand 20 nS transition times. TG1 is delayed by two 100 nS delays DL1 andDL2 to provide signals sequentially to momentarily close switches S8,S9, and S10, and to set BIS. TG2 is a descending ramp dT with a slope of1V/uS, representing the time remaining in the chopping cycle.

In order to obtain the a signal representing load current Ild withoutadding another input term, load is constructed from the current in L, asa running average over three cycles. The inductor current, representedby behavioral source B2, is integrated in C4, which is zeroed at thebeginning of each cycle by switch S10, thereby obtaining the totalcurrent for one cycle only, SIL. V6 controls switch S9 to sample andhold the previous cycle's SIL in C7 as midSIL. In similar fashion, V5controls switch S8 to sample and hold the second to last cycle'scurrent, lastSIL, in C3. Note that in a digital implementation, thisfunction can be provided by simply storing values in registers.Behavioral source BSIL averages the three inductor current integralsstored in C4, C7 and C3. The average is further smoothed with an analogfilter comprising of R2 and C5 to produce the term avgSIL. In order topredict the future load, it is necessary to introduce time. Behavioralmultiplier BMIL multiplies SILavg by the dT time remaining in the cycleby the output voltage, VO, to obtain the predicted load energy, KEload.BMIL also multiplies these terms by a constant signal 1.5 to enlargepredicted load energy by 50% to adjust KEload for the conducted energythat is not explicitly accounted for in the equations underlying thisembodiment.

The capacitive energy signal KECm comprises a required-capacitive-changesignal, dCapK, which is the difference of the squares of a modifiedinput voltage VI and the output voltage VO. To modify VI, behavioralsummer BSVI adds VI to its own average Vlavg and subtracts therefrom anaverage of VO, VOavg to generate a modified input signal VIm. Behavioralmultiplier BM1 then squares VIm to generate the signal VIm^2. Behavioralmultiplier BM2 squares VO to generate the signal VO^2.

A behavioral sourced B3 generates a signal VC responsive the value ofCI/2.5. VC may simply be a fixed representation of a fixed value of CI.Behavioral multiplier BMB multiplies CapK by VC, generating thecapacitive energy signal KECm, corresponding to the term KEC of FIG. 10.

In this case, KECm comprises, rather than a simple difference of thesquares of VI and VO, an adaptive term modifying VI. Thus conditioningVI brings the average output, VOavg closer to the average input Vlavgthan would occur using an unmodified VI. The main advantage of theadaptive term is that external load capacitance can be added dynamicallywithout causing loss of regulation or DC accuracy. Relatively slowchanges of the value of the switched inductor due to temperature orother operating conditions are also absorbed by the adaptive term.

To provide the rest of the classic capacitive energy equation KE=½C*E^2,the dCapK capacitive energy term would otherwise be multiplied by thecapacitance divided by 2. In this case, in the behavioral source B3 wedivide by 2.5 instead to obtain VC. Thus diminishing the capacitiveenergy term KECm reduces its error correction gain, thereby increase thesampled loop stability. This adaptive feedback loop makes up any smallDC error generated in its operation.

Behavioral summer BS4 compares the inductive energy signal KEL with thesum of a load-energy responsive signal KEld and the capacitive-energysignal KECm. When the inductive energy is greater than or equal to thesum of the capacitive energy difference and the predicted load energy,behavioral summer BS4 generates the balance energy signal BAL whichresets the flip-flop BIS, thus opening S5 and terminating the energizingof L.

The cycle repeats at the next set pulse S, regardless of whether theinductor has had time to fully be de-energized. Because any energy leftin the inductor is correctly represented by the KEL term, this amplifierpasses smoothly in and out of continuous mode, CCM, without distortionor excess ripple, even, as shown, while driving 200 uf at the output andbeing loaded with a sinusoidal current varying from 300 to 900 mA.Conventional converters tend to alternately under- and over-shoot at thevolatile boundary of continuous mode, generating subharmonic ripple,perhaps causing oscillation, or even generating positive feedback, withits consequent inclination toward self-destruction.

FIG. 13 shows the performance of the circuit in FIG. 12 with theadaptive correction applied in phase. VO is the output, tracking VI, theinput. The two traces are overlapping and practically indistinguishable.The lower trace, ILoad is the load current. Note that the only visibleeffect of increased load current is a slight increase in output ripple.The results shown in FIGS. 13-15 are based on SPICE simulations as areall other results set forth in this application unless otherwiseindicated. SPICE is a well known and commonly used tool for electricalcircuit simulation.

FIG. 14 is an enlarged detail of the input and output traces from FIG.13, plus the current IL in the switched inductor, L. IL passes out of,then back into, continuous conduction with no disruption of the outputvoltage. Continuous conduction periods are labeled CCM, thediscontinuous period is labeled DCM.

FIG. 15 is a detail of the input and output traces showing the responseto a change in load from 300 to 600 mA and back. Rise and fall times are1 uS for the load change. It is evident that the filter pole created forin-phase adaptive correction does not significantly impair the excellenttransient response that is made possible by the predictive energybalance. Non-predictive flyback converters would severely overshootduring the voltage downslope when the load was instantaneously cut inhalf. Not having means to actively correct the overshoot, theconventional converter would take an excessive amount of time to recoverfrom that transient.

Example 2 Recirculating Inductor Current for Improved Regulation inContinuous Mode DC-DC Converters

FIG. 16 depicts a flyback power converter with circulation of inductivecurrent according to this disclosure (State VI of FIG. 11F).

A fundamental problem in controlling CCM power converters occurs whenthe energy in the inductor L is not needed to supply the load current,since there is normally no place for that energy to go except to theload LD. The result is overshoot at the output when the load current isreduced. Because the continuous energy stored in the switched inductor Lis often large in magnitude compared to the energy moved from the inputto the output in any one switching cycle, a better way to handle thatsurplus energy can result in significant regulation improvements.Bidirectional converters can return the energy to the input, thoughextra switches are needed for the purpose (see example 4 below). Analternative is to recirculate the continuous current in the switchedinductor by closing one additional switch S7 connected directly acrossthe switched inductive element L. While the energy so circulated will besubject to resistive losses, most of the energy can be stored for shortperiods of time. In situations where the minimum load current is notnear zero, the recirculated energy can then be delivered to the loadgradually over a number of later cycles, as required. This arrangementallows high power density, good regulation and good efficiency in asimple system. The recent development of very low on-resistance FETswitches M2, M3 has made this approach practical.

Referring to FIG. 16, there appears outside the box labeled RECIRC mostof a conventional flyback power converter comprising and inductor, L, aninductive energy sensor, SENS, a switch, S5, a diode, S6, a filtercapacitor, CI, a snubber capacitor, C8, a snubber resistor, R26, avoltage output, VO, a input voltage source, V producing an inputvoltage, VI, a voltage comparator, CVO for comparing VO to VI, an S-Rflip-flop, BIS for turning on and off S5, a clock, CLK for periodicallysetting BIS, a limit threshold source, VLIM, and an inductive currentcomparator, COC for resetting BIS to turn off S5 should inductivecurrent IL exceed a threshold set by VLIM. What is absent, outside thebox labeled RECIRC in FIG. 16, from a prior art converter is a feedbackloop to reset BIS to attempt to control VO. An energy source P applies avoltage to power line VP to power the converter, and a load LD consumesconverter output power. The inventors have found that the FET switch S5implemented with the Philips part number PH2625, and the diode S6 isimplemented with the common 1N5817 work well. The operation of theseprior art components is, save the feedback loop according to thisdisclosure, conventional and well-known in the art.

One may note in FIG. 16 the absence of the usual OR-gate for resettingBIS to limit S5 ON-time, and thus control inductive current IL. Thisabsence occurs because this converter does not reset BIS as part of itsfeedback when VO exceeds VI. Instead CVO drives an AND-gate, RAND togenerate a recirculate signal REC which turns on the switch S7,initiating State VI of FIG. 11F. Had the converter just been in State IIof FIG. 11B, with current flowing in L, the energy thereof would ceaseto be transferred to load LD, and would begin to recirculated through S7as shown by the arrow labeled IREC. Were the circuit through S7 and Llossless, current would circulate therein without diminution for anindefinite time. However both S7 and L have series resistance causingcurrent to decay some percentage per unit time in according theinductive time-constant T=L/R where: T is the time in seconds forcurrent to decay 63.2%, L is the inductance in Henries and, R is theresistance of the circuit in Ohms. If the decay is relatively fast,inductive eddy current losses may also occur. In the converter of FIG.16, the peak inductive current during a recicrculating chopping-cycle isset by VLIM. If the load uses energy, inductive current will bediminished, but at the next clock signal instead of having to rampcurrent from zero, current may be ramped from some positive value tothat set by VLIM, substantially restoring to the reservoir ofrecirculating inductive energy that amount of energy having beenconsumed by the load and by losses.

Referring to S7 of FIG. 16, signal REC drives a FET driver DRV ofconventional character to turn on bidirectionally blocking andconducting switch FETs M2 and M3, also preferably like Philips partnumber PH2625. To cause S7 to conduct, M2 and M3 must be enhanced by DRVwith a voltage higher than VP. To that end, an auxiliary voltage sourceVTO, equal to the turn-on voltage of M2 and M3 is stacked upon VP andmay be connected to DRV by connecting nodes A and C. Alternatively, ifVO is never required to be less than VP plus VTO, the source VTO may beomitted and replaced by VO by connecting nodes A and B. An auxiliarytype-D flip-flop, BISR, is set by the clock and reset through gate RANDto terminate recirculation, thus initiating FIG. 11B State II should VOfall below VI. Recirculation is inhibited by BIS /Q during FIG. 11AState I through gate RAND.

FIG. 17 shows a converter employing recirculation according to thisdisclosure as taught in FIG. 16 combined with the predictive energybalancing techniques of this disclosure. Whilst the converter of FIG. 16provides good regulation without energy balancing techniques, it does soby incurring the efficiency loss of maintaining a constant reservoir ofcirculating inductive current. In situations where operationalconditions are fairly constant, the efficiency penalty caused by thepreset current can be small. The converter of FIG. 17 embodies, inaddition to recirculation according to this disclosure, predictiveenergy balancing according to this disclosure to minimize the amount ofcurrent that must be recirculated to provide good transient response.

The recirculation flyback example of FIG. 17 uses an approximate formulato determine the current required in L to provide a slight surplus ofenergy at the output side. The small surplus is preserved byrecirculation in the switched inductor. In addition, an adaptivefeedback path is provided to fine-tune the balance formula to end theenergize mode such that recirculation is needed for only a brief periodduring each cycle. For simplicity, the recirculation principle isapplied here to a unidirectional flyback converter using one switch S5and one diode S6, plus the bipolar recirculating switch S7, comprisingof M2 and M3.

The adaptive feedback methods described here could be used to adaptivelyadjust the preset current limit to achieve an appropriate recirculationtime, that time being the interval during which the recirculation switchis closed.

The portion of FIG. 17 lying outside the box labeled RECIRC-ADAPTcomprises a simple, prior-art flyback converter just as in FIG. 16. Thecomponents used for this embodiment are: R3 5K, R26 150 ohms, C2 100 nF,C8 820 pF, CI 330 uF, L 5 uH, FET switches S5, S7 all PH2625, and S61N5817.

As in FIG. 16, the prior art feedback loop is incomplete, creating aneed that is fulfilled by an improved feedback loop according to thisdisclosure described below. Unlike FIG. 16, this converter, like theprior art, comprises a reset OR-gate ROR, to provide an extra input forresetting BIS when a feedback criterion has been met. Unlike the priorart, this input responds to energy prediction rather than history.

Behavioral calculator B1 implements the following formula to implementenergy prediction:BAL is true if: (IL>VI*Ild*0.45+VO/VP*1.4+(VI−VO)*15+(1−FB)*0.15.

The L current needed is proportional to the weighted sum of three terms:the product of VI times the load LD current, the ratio of the VO to VP,and the surplus or deficit of voltage VO at the output capacitor CI. Inaddition, an adaptive feedback term, FB is also included in thecalculation of behavioral prediction calculator B1. FB is scaled andadded to the total to account for changes in either load capacitance orswitched inductance, as well as for thermal effects and any otherunquantified variables. The feedback signal, FB, is the duty cycle ofthe recirculation control signal, averaged by R3 and C2. The calculationof B1, for determining when to switch from energize to transfer modeadaptively attempts to minimize the recirculation time based on therecent history as stored in C2. By this means, typical recirculationtime is kept short, improving overall efficiency.

Edge-triggered bistable BISR and AND-gate RAND control recirculation.BISR is set by the synchronous clock to assure that an opportunity torecirculate inductor current is available only once per cycle. If theoutput voltage, VO, crosses above the input voltage, VI, during theflyback period, RAND begins a recirculation period by activating S7. IfVO falls below VI, BISR is cleared by a positive edge on the clockterminal. Otherwise, recirculation will continue until the end of theflyback period. AND-gate RAND prevents simultaneous conduction byforcing the recirculating switch to be off when switch S5 is on.

To provide the current sensing both of IL and of Ild behavioralsimulation blocks were used. For slower converters, these functions canbe embodied by well-known current monitors, but for high speedconverters either high speed amplifiers or composites ofcurrent-transformers supplemented by amplifiers to provide DC paths, asis taught in application Ser. No. 11/593,702, here incorporated byreference, may be used.

The principle of recirculation described here is not limited to flybackconverters. It applies equally to buck and combination topologies. Inall cases, recirculation can ease the difficulties of controlling CCMpower converters.

V provides the input voltage VI. For a DC-to-DC converter, VI wouldnormally be a constant voltage.

FIG. 18 shows SPICE-generated waveforms of the circuit of FIG. 17. InFIG. 18, VI is a 2 volt peak-to-peak sine wave centered at 5 volts, sothat circuit behaves as a self-powered amplifier. FIG. 18 shows VP to bea sine wave centered on 2.5 volts and of 0.5 volts peak-to-peakamplitude. The load current, Ild, ramps from 0.4 to 4 amps and back. Theswitched inductor current, IL, is seen to be continuous for the entiretime. The VI and VO traces superimpose, and are indistinguishable inthis figure, with more ripple detectable during the period of higherload.

FIG. 19 is a detail of FIG. 18 showing the recirculating action duringthe negative load transient. Note the substantially flat areas on theinductor current waveform, IL, corresponding to periods ofrecirculation, and the absence of overshoot as the load drops.

Note that if S6 acts as a synchronous rectifier during transfer, and isopen during recirculation instead of acting as a diode duringrecirculation, that this converter is capable of stepdown, as well asstepup, operation.

Example 3 Bidirectional Flyback Amplifier with Energy Balance andContinuous Mode Operation

Additional considerations must be observed when performing energybalance control for continuous conduction mode (CCM) power converterscompared to discontinuous mode (DCM) converters. This example, FIGS. 20Aand 20B, shows a bidirectional flyback amplifier with CCM and DCM energybalance well under control. This embodiment provides excellentefficiency because it returns any excess energy at the load to thesource. Because this device functions as its own step-up power supply,the inefficiencies of conventional regulated power supplies aresubstantially avoided. The simulated load current shown is bipolar andsymmetrical around zero. Under these circumstances, over one load cyclethe net power consumption is near zero because energy is alternatelytransferred to and recovered from the output reactance. Also, thistopology and control method performs exceptionally well as a powersupply when a DC reference is provided in place of the AC referencesignal.

Referring to FIG. 20A, four switches, S2, S3, S5 and S6 perform StatesI, II, III, and IV of FIGS. 11A-11D. State I is forward flybackenergize, State II is forward flyback transfer, State III is reverseflyback energize and State IV is a reverse flyback transfer.

These states are driven by three type-D flip-flops or bistables, BISM,BISR, and BISF. All three bistables are clocked by a rising edge CLK,having a 750 nS true time every 5 uS, to establish a chopping cycle.

Upon the rising clock and in accordance with the state of thereverse-balance signal RBL upon its D-input, BISM generates a modesignal M that will persist for the balance of the chopping cycle beingcommenced.

The RBL signal is generated by comparator CKEC which reports the sign ofthe capacitive error energy dCapK, as shown in FIG. 20B.

If RBL is true, M will be true and the cycle being commenced willcomprise, in succession, the forward States I and II to effect a forwardflyback energy transfer from PA to PB. State I first occurs to energizeL from PA. Due to the minimum true-time of CLK and its connection toBISF, PRE, the minimum time for energizing L will be substantially theminimum true-time. After the minimum true-time, for the balance of thechopping cycle, State I may be terminated and State II initiatedresponsive the forward balance signal FBL, which indicates that theforward balance calculator described below has predictively determinedthat L contains sufficient energy from source P to supply the energydemands of PB. Upon such determination, FBL asynchronously clears BISF,initiating State II, which persists until the chopping cycle ends.

If RBL is false, M will be false and the cycle being commenced willcomprise, in succession, the reverse States III and IV to effect areverse flyback energy transfer from PB to PA. State III first occurs toenergize L from PB. State III may be terminated and State IV beinitiated responsive the forward balance signal RBL, which indicates thereverse balance calculator described below has non-predictivelydetermined that L has removed sufficient energy from PB, whereupon RBLasynchronously sets BISR, initiating State IV, which persists until thechopping cycle ends.

Between the aforementioned bistables and switches is a decodercomprising AND-gate ANDI for generating a signal I corresponding toState I of FIG. 11, and AND-gates ANDII, ANDIII, and ANDIV for likewisegenerating their respectively corresponding signals II, III, and IV.This decoder may be embodied as discrete logic as shown, as a behavioralstate decoder as will be shown in other figures, as a well-known PLD, asa table executed from a processor, or by other well-known means.

Through OR-gate OR5, switch S5 functions unconditionally as abidirectional switch in State I. However also through OR5 and throughAND-gate SRAND5, and responsive to synchronous rectifier comparatorSRC5, S5 also acts as a synchronous rectifier during State IV. Here, asin the International Rectifier part IR1167, synchronous rectification isbe based on switch voltage, but may alternatively be based on switchcurrent, as is well-known. The operation of S6, OR6, SRAND6, and SRC6 isanalogous to that of S5 and its associated circuitry.

As taught above, and also well-known, an inductive energy sensor SENSgenerates a signal VIL responsive to current, field or E-T product in L.

This converter may comprise inductive current limiting and outputvoltage limiting described above and also well-known.

Referring now to FIG. 20B, there is shown the timing and balancingcircuitry that generates the signal CLK, RBL, and FBL that control thebistables of FIG. 20A.

Timing generator TG generates a clock signal CLK, a pre-clock signalPCK, and a descending ramp dT, as taught in FIG. 12, representing timeremaining in a chopping cycle. Here CLK has a true-time that is used toset a minimum pulse width. PCK actually occurs just before the end of aprevious cycle. Comparator CTM compares dT with a threshold Tmax to setminimum forward transfer time, based on the remaining time in thechopping cycle, overriding through OR-gate BALOR the balance signal BALand initiating State II of FIG. 11B.

In this simulation, TG provides a synchronous 750 nS CLK pulse every 5us. TG also supplies late in a previous cycle, an identical pulse, PCK,50 ns prior to the CLK pulse closes switch S11 to store the previouscycle's KEL value in C12. TG generates dT with a slope of −1V/uS.

The RBL signal is generated by comparator CKEC which reports the sign ofthe capacitive energy difference dCapK.

KEL is the inductive energy calculator described in FIG. 12. Since, inthis converter, inductive energy can be of either polarity KEL's outputis processed by multiplier MSGN and comparator CILSGN to endow it with asign responsive to the direction of inductive current to generate asignal KELs.

At the end of a previous cycle PCK briefly closes S11 to store in C12 asample of the “pedestal” KEL of IL, which relates to the energyremaining in L at cycle's end, Klastped. Through pedestal summer SUMPEDthe predicted pedestal Kpp is subtracted from Klastped, and scaled bythe factor 0.9 in multiplier MPED, this predicted change in pedestal issubtracted from KELs to generate a new inductive energy signal, avKEL.

A predictive pedestal current calculator comprising S12, CPRRCT, SUMdVL,S13, CILdMIN, DIVdIL, MdIL, and SUMILd, generates a signal PREDICTpredicting the current at present cycle's end. The signal PREDICT isgenerated as follows: Summer SUMdVL generates a signal dVL representingthe de-energizing voltage across L during State II. MdIL multiplies dVLby dT, representing remaining cycle time, to generate a signal which isdivided by VL, a voltage representing the value of L as taught in FIG.12, by divider DIVdIL to generate a signal dIL, the availablede-energization of inductive current. Signal dIL is computed accordingto the classic formula dI=E*dT/L where dI is current change, E isinductor voltage, and L is inductance.

SUMILd subtracts that predicted change from the present inductivecurrent, represented by VIL, to generate a signal, ILd, predictingcurrent at cycle's end. Comparator CILdMIN and switch S13 select thelesser of ILd and the constant 20 to prevent instability shouldimpossibly large pedestals be temporarily predicted.

Comparator CPRRCT and S12 half-wave precision-rectify the signalselected by S13 to generate the signal PREDICT. To convert PREDICT froma current signal to an energy-proportional signal it is squared andmultiplied by VL in MKpp to generate the predicted pedestal energysignal Kpp.

The avKEL calculation helps account for a seeming contradiction whichhas confounded prior art continuous control methods. Because inductiveenergy relates to the square of the current, more energy per ampere ofcurrent change can be transferred at higher continuous currents. If allthe energy in a CCM switched inductor is transferred to the load, theremay not be time to recharge the inductor to the level necessary duringthe next energize cycle. In order to avoid subharmonic behavior, thecontinuous inductive energy must be conserved for subsequent cycles. Forcontrol according to this disclosure, the energy to be retained in theinductor is mathematically subtracted from the total inductive energy,KEL, as is done in SUMKEL, term to obtain the signal avKEL, or availableinductive energy.

If the pedestal is predicted to decrease, the avKEL term is reduced,extending the energize period and thereby preserving more pedestalcurrent. Conversely, if the pedestal is predicted to increase, the avKELterm is augmented, causing earlier termination of the energize mode anda smaller increase in the pedestal current. This process has the effectof limiting targeted changes in pedestal energy to 10% per choppingcycle. Because of energize and transfer time limits, and becauseoperating conditions can change dynamically, actual cycle-to-cyclepedestal variation can be larger than the target percentage. The exactpercentage used in this calculation is non-critical. Even 100% producesacceptable results.

By using avKEL as the inductive energy term in the balance detected byCBAL, smooth behavior around the transition between DCM and CCM can beachieved. Any mechanism which has the effect of limiting thecycle-to-cycle variation of continuous current for improving regulationis practicing this disclosure.

The capacitive side of the balance equation is similar to the otherpredictive flyback converters of this disclosure. dCapK is thedifference in capacitive energy as described in FIG. 12. It ismultiplied by the capacitance by MKEC and compared with inductive energyby CBAL to generate the balance signal BAL.

Note that both DCM and CCM transfers are possible in both forward andreverse directions.

The capacitive term VCsc would yield unity loop gain with 0.5V perFarad. In this simulation it is slightly attenuated to 0.35 to reducesampled loop gain and thereby obtain better loop stability.

Adaptive methods can be used to account for changes in L and C, similarto those shown in FIG. 14. A load term could be added to speed responseto sudden load changes, also illustrated in FIG. 14.

FIG. 21 shows VP to be a sine wave centered on 3 volts and of 1 voltpeak-to-peak amplitude. The load current is a 1.5 KHz sine wave centeredon zero, swinging plus and minus 0.5 amps (not shown). The switchedinductor current, IL is seen to be both continuous and discontinuous inboth the positive and negative directions. The VI trace swings 10 voltspeak-to-peak at 1 kHz, centered on 14 volts. The output, VO, closelyfollows the VI signal. DCM and CCM periods are marked appropriately.

Example 4 Bidirectional Flyback Amplifier with Inductive Energy Storagefor Improved Agility

In situations where transient response is of paramount importance, extraenergy can be stored in the switched inductor so that it is immediatelyavailable to supply the load. The energy balancing techniques of thisdisclosure have been shown to be valuable for controlling converterswhich maintain extra energy in the inductor (see FIG. 10). Therecirculation technique, FIG. 18, is another means of storing extraenergy inductively. The five-switch example shown here in FIGS. 22through 24 is another novel way to approach the same problem. Again, forsimplicity, this example illustrates the new technique in a converterwithout energy balance. Energy balance could be added for improvedefficiency. In this case, regulation is near ideal, so little transientperformance could be gained from energy balancing.

This example shows a bidirectional flyback amplifier with bipolar outputwith any continuous conduction under control. The following advantagesof example 3 pertain here, as well: Because this device functions as itsown power regulator, the inefficiencies of conventional regulated powerconverters are avoided. The simulated load shown in FIG. 25 is bipolarand symmetrical around zero. Under these circumstances, the net powerconsumption for a complete load cycle is near zero because energy isalternately transferred to and recovered from the output. Efficiencieswell over 90% are practical using this approach. Also, this topology andcontrol method performs exceptionally well as a power supply when a DCinput is provided in place of the AC input signal VI as shown in FIGS.26 and 27.

Referring to FIG. 22, there is shown a state decoder STDEC forgenerating the States I through VIII of FIGS. 11A-11H responsive to theoutputs of flip-flops to be shown below in FIG. 24 in accordance withthe following logical expressions:State I=M&FState II=M&/F&/XState III=/M)&R&/FNState IV=/M)&/R)&/FNI/M&/R&FN&/XNState V=M&/F&XState VI=absent, used only in FIGS. 16 and 17State VII=/M&/R&FN&XNState VIII=/M&R&FNwhere the signals cited correspond to those illustrated in FIGS. 22 and23, and to the like-named States of FIGS. 11A-11H, and

“&” represents the logical AND function,

“|” represents the logical OR function,

the prefix “/” performs the logical NOT upon its suffix.

In this example shown more fully in FIG. 24 below, one state decoderSTDEC serves several switch-blocks, an example of which is illustratedin the switch-block SWBn of FIG. 23. It is to be understood that the nof SWBn is to be a digit, here 2 through 6, representing an instance ofSWBn corresponding to S1 through S6 of FIGS. 11A-11H.

The signals cited and present constitute a State bus STB of signals towhich SWBn responds. Each instance SWBn has a unique decode functionimplemented by switch decoder SWDEC, which is responsive to signals ofSTB and to a signal IS<Ith which is generated by a comparator CSWresponsive to voltage across switch S. Thus the SWBn performs the “untilIth” function. Decoder SWDEC generates a signal DRV to drive S to closea circuit between terminals SWm and SWn, the latter corresponding to theterminals of one switch of FIGS. 11A-11H. Decoder SWDEC responds in itsinstances to STB and IS<Ith in accordance with the following logicalexpressions:SWB2=(I&IS<Ith)|II|IVSWB3=((III|VII)&IS<Ith)|VSWB4=V&IS<IthSWB5=I|(IV&IS<Ith)SWB6=(II&IS<Ith)|III|VII

Referring to FIG. 24, five instances of SWBn, labeled SWB2 through SWB6,perform switch decoding and switching correctly to function as S2through S6 respectively of FIGS. 11A-11H.

The five-switch power converter of FIG. 24 is similar in topology, butnot in control, to the four-switch continuous example of FIGS. 20A and20B. S7 and State VI of FIG. 11F are not used in this example.

As in FIGS. 20A and 20B, a timing generator TG provides a set-pulse S tocommence each chopping cycle, causing BISM, BISF, and BISR, throughcomparator CVO, to latch data responsive to the polarity of anydifference between VO and VI. BISM holds that data until cycle-end.BISF, and BISR respond to asynchronous events as previously described inFIGS. 20A and 20B. SENS senses inductive energy as in FIGS. 20A and 20B.Thus, for moving positive energy from PA to PB, energize State I begins,followed by forward transfer State II. TG of FIG. 24 also comprises afast clock, CKF, that generates a plurality of transitions within aperiod of S. FIG. 24 further comprises a new flip-flop BISX whichfollows /M at the first clocking transition of CKF following S, thusresetting BISX early in a positive energy moving cycle. In FIGS. 20A and20B, State II would persist until cycle-end, but in FIG. 24 if theregulation voltage is reached, comparator CVO, through inverter INV,resets flip-flop BISX to invoke State V of FIG. 11E. When State Vsupersedes State II, returning excess inductive energy to PA rather thancreating excess voltage at PB.

FIG. 24 further comprises added switch S4, comprised by a switch-block,SWB4, to embody an additional transfer mode corresponding to State Vwhich returns un-transferred energy directly back to the power inputport. In FIG. 24, instead of precisely metering the energizing of Lduring the energize state, L is loaded with a substantial excess ofenergy controlled by inductive current comparators CP and CN responsiveto the difference between VIL and their respective limit signals LMP andLMN.

Thus, as in the six-switch example of this disclosure, the transferportion of the cycle is divided into sub-cycles, but with CKF running ata higher frequency than S there can be numerous sub-cycles per period ofS.

For the positive polarity, States II and V alternate, responsive to thecomparator CVO, for the remainder of the transfer time until the end ofthe chopping cycle. The inductor current continues to flow in the samedirection during State V, but is now directed back to the input port, asshown in FIG. 25. For the negative polarity, States VII and IV likewisealternate.

Such division of each cycle into three or more active portions (plus apossible additional period of depleted inductor current) for the purposeof improved regulation is unknown in the prior art. FIG. 25 shows theimprovement achieved by adding a 800 kHz clock for CKF. It is evidentfrom FIG. 25 that a multi-phase converter could reduce ripple bymaintaining one phase in transfer alternation at all times. The possibleimprovement in ripple reduction is more than a factor of four.

Reverse transfers here are essentially identical to the four-switchexample shown in FIGS. 20A and 20B.

In order to control negative voltages with either polarity of load, anew flip-flop, BISFN, which is set when the alternation of States IIIand IV are not sufficient to drive the output far enough negative. Underthose circumstances, AND-gate ANDAR, responsive to VO, to /M and,through CTH, the polarity of inductive current, will generate a clock toset BISFN. ANDAR determines that the inductor could not gain significantnegative charge during State III of a reverse transfer. BISFN, once set,causes State VIII to be used as the negative energize state, followed bythe State VII to transfer negative energy from the input to the output.

If sufficiency is obtained, bistable BISXN is cleared by the output ofcomparator CVO. That causes State IV to be invoked, to return excessnegative energy to the input. The bistable BISXN is clocked by CKF toprevent rapid alternation of States VII and IV. This negative forwardtransfer sequence matches the normal forward transfer sequence in allregards except the polarity of the inductor current.

This example uses no prediction to determine the appropriate duration ofthe forward or reverse energize modes. For regulation, it reliesentirely on returning excess energy to PA, and in so doing incurs asurprisingly small efficiency penalty. Small additional complexity wouldallow predictive or adaptive forward energize modes. That would permitan adaptive tradeoff between maximal efficiency and best transientresponse by adjusting the percentage of inductively stored energy. NoA/D converters save single-bit comparator functions are needed for adigital implementation of this technique and no computations morecomplex than magnitude comparisons, analog or digital, are required.

Note that the ratio of power input to output voltage in this example isnot constrained to a narrow range as in prior art devices. Thisadvantage is gained by avoiding PWM control. Also note that, with someadditional control logic, PA could be made bipolar without the additionof another switch because the inductor can be energized in eitherpolarity for the forward energize and can be discharged in eitherpolarity for the reverse transfer modes. Only the control logic toexchange S2 for S4 and S3 for S5 in the presence of a negative VP wouldbe needed.

A comparator that provides the polarity of the error voltage CVO and anovercurrent detector for the switched inductor, CP, CN, and ORLIM, arethe only required inputs if the switches have the “until Ith” functionbuilt-in, like the International Rectifier IRF 1167. The various switchcombinations can be programmed into to a solid state memory and indexedby address lines related to time, polarity and overcurrent as in thewell known “state machine”, can be programmed into a microcontroller, orother programmable logic device, or can be reproduced using discretecomponents.

Note that though this example is based on additive flyback forwardtransfers, it can faithfully reproduce output voltages that are belowthe input voltage and are positive or negative. The ability to maintaingood regulation for step-up and step-down operation with as few as twocomparators for control is unique to this disclosure.

FIG. 25 shows the behavior with a 1 kHz sine wave input signal, VI, witha peak-to-peak amplitude of 100 volts, centered on zero volts. The loadcurrent switches from plus 2 A to minus 2 A at 350 uS, back to plus 2amps at 700 uS, and then to minus 2 A at 1050 uS. The power inputvoltage, VP, is a 2 kHz sine wave varying from 3 to 157 volts. Theoutput voltage VO and input voltage VI are superimposed and are hard todistinguish here. The inductor current, IL, is seen to be bothcontinuous and discontinuous in both the positive and negativepolarities. Note the very minor output disruption at the loadtransitions due to continuous mode operation. Load transients that donot cross the DCM/CCM boundary are nearly invisible. The ability of anadditive flyback converter to handle output voltages much above, muchbelow, or equal to the power input voltage is novel.

FIG. 26 shows multiple states being used during single chopping cycles.The input voltage VI is steady at 6 volts. At 1.002 mS, the load changesfrom minus 2 to plus 2 amps. Before the load transition, States III andIV are alternating to effect reverse energizing and transfer. After theload transition, the inductor current moves in the positive directionduring the forward energize, State I. It rises for approximately 2 uS,then more slowly (due to the additive nature of the transfer) during thebrief forward transfer, State II. State V follows, causing inductorcurrent to flow back to the input PA. States II and V alternate oncemore before the end of the chopping cycle. The next forward cycle beginswith State I at 1.01 mS. Note that the VO trace just touches the VItrace at the end of each asynchronously terminated state, except forState I, which is terminated by the inductive current limit.

FIG. 27 also shows the use of three states per chopping cycle, this timewith the input voltage VI steady at −6 volts. State VIII energizes theinductor, followed by State VII, which transfers energy untilsufficiency, causing State IV to be invoked. State IV returns excessenergy from the inductor to the input side. States VII and IV alternateonce more before the end of the chopping cycle.

In a flyback converter it is usual for a diode or a synchronousrectifier, corresponding to S6 of FIGS. 11A-11H, to switch ON upon thecondition that inductor voltage exceeds output voltage, VO. According tothis disclosure, adding to that basis the condition that desired inputvoltage, VI, exceed output voltage, VO, as shown in FIG. 24 allows thisflyback converter, unlike the prior art, to generate VO substantiallyless than the inductor energizing voltage VP. To obtain thisimprovement, should the magnitude of VO exceed that of VI during StateII, State V is invoked, turning off S6 to avoid generation of excess VO.It should be understood that an alternative path for any current in Lmust be provided to avoid excessive voltage. The recirculating examplesof FIGS. 16 and 17 provide such a path by recirculation. FIG. 24provides such a path by invoking State V after forward transfers andState IV after negative forward transfers.

In FIG. 24, this improvement is embodied through the term “X” thatappears in the logical definition of State V,State V=M&/F&X

Without adaptive techniques, recirculation according to FIG. 16 orreverse transfer according to FIG. 24 will incur some inefficiency.However if an adaptive technique such as that shown in FIG. 17 beemployed to change the current limit responsive to the time spent inrecirculation, or to minimize the time of energy return, efficiency maybe maximized.

Higher IL pedestal currents tend toward better transient response butmay adversely affect efficiency. The current limit may be set inadvance, or adaptively, to adjust the tradeoff between efficiency andtransient response.

Since the converters of FIGS. 16, 17, and 24 make only current limitdecisions during the forward energize state, extremely short energizetimes can be safely controlled. The result is stable operation withhigher VP voltages than are practical with conventional converters thattolerate only limited ratios of VP and VO.

VP must be high enough to energize the switched inductor sufficiently inthe energize time available, and lower than the voltage rating of theinput switches. Those limits are much broader than the existing art.

The combination of tolerating much higher input voltages, and theability to produce a VO below VP make flyback power converters accordingto this disclosure much more flexible than prior-art power converters. Aprior-art buck converter only produces voltages below VP, a prior-artboost or flyback converter normally requires VO to be greater than VP, abuck/boost inverter usually requires the absolute value of VO to begreater than VP, and a SEPIC converter allows VO to equal VP but at thecost of a second inductor. The buck/boost topology is the most flexiblefor input and output ranges but it is rarely seen. Lackluster efficiencyand difficulty controlling the buck/boost topology using PWM techniquesmay explain that rarity. Buck/boost converters, when functioning in buckmode, show instability near the DCM/CCM boundary. Also, the lack ofrecirculation or energy return capability often leads to the stabilityissues common to this type of converter, obviating much of the potentialadvantage of extended input and output range.

Conventional converters have additional constraints on the ratio of VOto VP due to the need to avoid extreme duty cycles. Accepting a VP rangeof more than five to one is rare in conventional converters and thosethat have extended input ranges are often limited to a single outputvoltage. Those conventional converters that do have adjustable outputsare normally only adjustable over a narrow range. Example 4 shows aflyback converter that can accept a VP varying by more than 50 to 1, andcan produce a VO from zero to more than ten times VP in either polarity.It can also allow VO to equal VP. Because that flexibility can be gainedwith only comparators and logic functions for control, power convertersbased on example 4 can economically replace many different narrow-rangepower converter products.

Example 5 Application to CCM Flyback Regulation

A fundamental difference exists between the discontinuous conductionmode (DCM) and continuous conduction mode (CCM) operation of a flybackconverter.

In the DCM, inductor current is reset to zero by the end of a choppingcycle. There is, therefore, no memory of previous cycles carried in theinductor.

Since inductive energy is proportional to the square of current, a givencurrent change can transfer more energy at a higher current than at alower current. Since current change is proportional to time, more energyper unit time, both into and out of the inductor, can be moved throughthe inductor at a higher current.

As long as a converter remains in the DCM mode, the current needed tosupply the load is attained within a given cycle, but in the CCM mode anon-zero inductor current. or pedestal, continues from cycle to cycle.When load changes, the pedestal must be adjusted. With a given inductor,higher loads require higher pedestals than lower loads. Since inductorcurrent cannot be changed instantaneously, time is required to adjustthe pedestal to sustain an increased load, and many cycles may berequired to increase or to decrease the pedestal. Managing the CCMflyback converter pedestal using restraint and the neededcounter-intuitive response is an aspect of this disclosure. There aremany ways to implement such management. Usually pedestal management forthe time-constrained synchronous converter is the difficult case, sothat will be addressed below.

In one management method according to this disclosure, shown in Example3, the pedestal that would remain at cycle end, should a transfer statebe immediately initiated, is predicted during the energize state basedon inductance, the voltage across the inductor, and the time remainingwithin the cycle for an energy transfer state. If the majority of theenergy represented by the predicted pedestal current is retained for thesubsequent cycle, both excellent regulation and well-damped response canbe obtained. This method is very helpful for load increases.

When load increases, output voltage tends to fall, causing prior artregulators to increase energize-mode duty-cycle, often so precipitouslythat the remaining cycle time is insufficient for efficient energytransfer to the load. Such a response exacerbates the transient decreaseof output voltage, and often incurs subsequent over-correction.Moderation according to this disclosure is practiced by predicting thepedestal value and limiting its per-cycle change to damp the transientresponse of the regulator.

Another management method uses a compound prediction. The maximum energydeliverable in the next cycle is based on the predicted pedestal at theend of the present cycle. If and when that maximum energy falls belowthe energy predicted to be delivered in the present cycle, the presentenergize mode is terminated. Note that prior art regulators respond topast cycles rather than to predictions of present and future cycles.

The present disclosure is practiced whenever predicted present cycle orfuture cycle pedestal currents, in combination with or ignoring pastpedestal currents, are conserved or reduced for future regulation. Thepresent disclosure is practiced whether the predictions used forpedestal control are responsive to inductor current, magnetic field, orcalculated integration of inductor voltage-time product.

Pedestal control according to this disclosure does not interfere withDCM converter operation, inasmuch as the pedestal simply becomes zeroand has no effect. Transitions between DCM and CCM are graceful as shownin FIG. 21.

Pedestal control moderates energy flow to minimize load-inducedtransient disturbances, but does not add or subtract another energysource substantially to eliminate such disturbances. Just as predictioncan be used to mitigate disturbances, is can also be used to connectenergy sources and/or sinks to the load or switch taps on the inductor.Such switching exploits the advanced knowledge gained by predictionsubstantially to cancel transients. An arrangement that is particularlyadvantageous in the common case of a load that alternates between twolevels is to switch any overshoot energy from a load decrease into astorage capacitor. When the load resumes its former level, the energy inthat capacitor is switched to cause it to support the load while thepedestal is readjusted. This disclosure is also practiced when pedestalprediction is used to connect an auxiliary energy source or sink to theinductor and/or load. In many converters, the switches required fortransient cancellation already exist and need merely to be correctlyconfigured to effect cancellation. Such prediction can also be used toinvoke inductive energy storage by circulation as taught elsewhere inthis application.

It is understood that the disclosure is not limited to the disclosedembodiments, but on the contrary, is intended to cover variousmodifications and equivalent arrangements included within the spirit andscope of the appended claims. Without further elaboration, the foregoingwill so fully illustrate the disclosure, that others may by current orfuture knowledge, readily adapt the same for use under the variousconditions of service.

We claim:
 1. A switched-mode power converter comprising: at least twopower-moving ports comprising a power input port and a power outputport; an inductive reactor; a plurality of switches configured toselectively connect the power input and output ports to the inductivereactor; demand computation circuitry configured to compute an energydemand for the converter based on (i) a reference signal and (ii) anoutput signal based on a voltage or current at the power output port;supply computation circuitry configured to compute an availableinductive energy supply based on instantaneous inductive energypresently stored in the inductive reactor, wherein the availableinductive energy supply is an amount of inductive energy presentlystored in the inductive reactor that is available to be transferred tothe power output port during a present chopping cycle for the converter;and control circuitry configured to control the switches based on (i)the energy demand and (ii) the available inductive energy supply.
 2. Theconverter of claim 1, wherein the supply computation circuitry isconfigured to compute the instantaneous inductive energy based oncurrent in an inductor winding.
 3. The converter of claim 1, wherein thesupply computation circuitry is configured to compute the instantaneousinductive energy based on a signal from a magnetic field sensor.
 4. Theconverter of claim 1, wherein the supply computation circuitry isconfigured to compute the instantaneous inductive energy based onintegration of a volt-time product having been applied to the inductivereactor.
 5. The converter of claim 1, wherein, if (a) the switches areconfigured to store input energy received at the power input port intothe inductive reactor and (b) the control circuitry determines that theavailable inductive energy supply reaches the energy demand, then thecontrol circuitry re-configures the switches to (i) stop storing theinput energy into the inductive reactor and (ii) start transferringenergy stored in the inductive reactor to the power output port.
 6. Theconverter of claim 1, wherein the demand computation circuitry isconfigured to compute the energy demand based on a difference betweeninstantaneous energy at the power output port and desired energy at thepower output port.
 7. The converter of claim 1, wherein the supplycomputation circuitry is configured to compute the available inductiveenergy supply based on (i) previous inductive energy at end of aprevious chopping cycle; (ii) predicted inductive energy at end of thepresent chopping cycle; and (iii) the instantaneous inductive energy. 8.The converter of claim 7, wherein: the supply computation circuitry isconfigured to compute the predicted inductive energy at the end of thepresent chopping cycle; and the control circuitry is configured tocontrol the switches so as to limit a per cycle change in inductiveenergy at the end of each chopping cycle.
 9. The converter of claim 7,wherein the supply computation circuitry is configured to compute theavailable inductive energy supply by: (a) computing an energy differencebetween (i) the previous inductive energy at the end of the previouschopping cycle and (ii) the predicted inductive energy at the end of thepresent chopping cycle; (b) computing a scaled energy difference basedon a product of (i) the energy difference and (ii) a scale factor havinga magnitude less than one; and (c) computing the available inductiveenergy supply based on a difference between (i) the scaled energydifference and (ii) the instantaneous inductive energy.
 10. Theconverter of claim 1, wherein the control circuitry is configured toregulate the output voltage or current from chopping cycle to choppingcycle while predictively managing the inductive energy at the end ofeach chopping cycle.
 11. A method of controlling a switched-mode powerconverter comprising a power input port, switches, an inductor, and apower output port, said method comprising: transferring energy from thepower input port to energize the inductor; computing an energy demandfor the converter based on (i) a reference signal and (ii) an outputsignal based on a voltage or current at the power output port; computingan available inductive energy supply based on instantaneous inductiveenergy presently stored in the inductor, wherein the available inductiveenergy supply is an amount of inductive energy presently stored in theinductor that is available to be transferred to the output port during apresent chopping cycle for the converter; and controlling the switchesto transfer energy from the inductor to the power output port based on(i) the energy demand and (ii) the available inductive energy supply.12. The method of claim 11, wherein the power converter operates incontinuous conduction mode.
 13. The method of claim 11, furthercomprising application of a fast clock to divide a basic chopping periodof the converter.
 14. The method of claim 11, wherein said converteroperates in any of bipolar, bidirectional, inverting, or flyback modes.15. The method of claim 14, wherein the converter is a flyback converterand voltage at the power output port is controlled to be below voltageat the power input port.
 16. The method of claim 11, wherein, if (a) theswitches are configured to store input energy received at the powerinput port into the inductor and (b) the available inductive energysupply is determined to reach the energy demand, then the switches arere-configured to (i) stop storing the input energy into the inductor and(ii) start transferring energy stored in the inductor to the poweroutput port.
 17. The method of claim 11, wherein the energy demand iscomputed based on a difference between instantaneous energy at the poweroutput port and desired energy at the power output port.
 18. The methodof claim 11, wherein the available inductive energy supply is computedbased on (i) previous inductive energy at end of a previous choppingcycle; (ii) predicted inductive energy at end of the present choppingcycle; and (iii) the instantaneous inductive energy.
 19. The method ofclaim 18, wherein the available inductive energy supply is computed by:(a) computing an energy difference between (i) the previous inductiveenergy at the end of the previous chopping cycle and (ii) the predictedinductive energy at the end of the present chopping cycle; (b) computinga scaled energy difference based on a product of (i) the energydifference and (ii) a scale factor having a magnitude less than one; and(c) computing the available inductive energy supply based on adifference between (i) the scaled energy difference and (ii) theinstantaneous inductive energy.
 20. The method of claim 11, wherein theswitches are controlled to regulate the output voltage or current fromchopping cycle to chopping cycle while predictively managing theinductive energy at the end of each chopping cycle.